Phase error detecting device, waveform shaping device and optical disc device

ABSTRACT

A waveform shaping portion receives a digital reproduced signal generated from an analog reproduced signal reproduced from an information recording medium and shapes the waveform of the digital reproduced signal. A maximum likelihood decoding portion applies maximum likelihood decoding to the digital reproduced signal in the shaped waveform and generates a binarized signal indicating the result of the maximum likelihood decoding. A phase detection portion extracts, during the maximum likelihood decoding, a phase error using state transition patterns having only a single zero cross point among differential metrics at a plurality of merging points at which a set of paths branched from a given state merges. A synchronization detection portion generates a reproduction clock signal using the phase error that has been detected and brings the digital reproduced signal into synchronization with the reproduction clock signal that has been generated. This configuration makes it possible to generate a reproduction clock signal in a stable manner.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to a phase error detecting device, awaveform shaping device, and an optical disc device for reproducing asignal recorded over the limit of optical resolution, and moreparticularly, to a phase error detecting device, a waveform shapingdevice, and an optical disc device for performing phase synchronizationon a reproduced signal using maximum likelihood decoding, such asViterbi decoding.

2. Description of the Background Art

Recently, the shortest mark length in recording is nearing the limit ofoptical resolution because of the density of an optical disc that isbecoming higher and higher. An increase in intersymbol interference anddeterioration of SNR (Signal Noise Rate) are therefore becoming morenoticeable. Under these circumstances, it is becoming popular to adoptPRML (Partial Response Maximum Likelihood) as a signal processingmethod. PRML is a technique combining partial response (PR) and maximumlikelihood decoding (ML). It is a method for selecting a most likelysignal sequence from a reproduced waveform based on the premise thatknown intersymbol interference will occur. This method is thereforeknown to achieve better performance than a conventional leveldetermination method.

Meanwhile, a shift of the signal processing method from the leveldetermination method to PRML poses a problem as to an evaluation methodof a reproduced signal. Jitter, which is a reproduced signal evaluationindex that has been used conventionally, is premised on the signalprocessing by the level evaluation method. Accordingly, there are caseswhere jitter is not correlated with the performance of PRML that uses adifferent signal processing algorithm from the one used in the leveldetermination method. Under these circumstances, various new indicescorrelated with the performance of PRML have been proposed (see, forexample, JP-A-2003-141823).

Also, a new index that enables a detection of displacement (edgeshifting) between marks and spaces, which is crucial to the recordingquality of an optical disc, is now being proposed. When PRML is adopted,this index should also be in conformity with the concept of PRML,correlated with the performance of PRML, and able to indicate adirection and an amount of edge shifting quantitatively pattern bypattern (see, for example, JP-A-2004-335079).

In addition, as an optical disc becomes further denser, intersymbolinterference and deterioration of SNR will increase. In this case, it ispossible to maintain the system margin by adopting high-order PRML. Forexample, an optical disc having a diameter of 12 cm and a recordingcapacity of 25 GB per layer is able to maintain the system margin byadopting PR 1221 ML. However, it is necessary to adopt PR 12221 ML foran optical disc having a recording capacity of 33.3 GB per layer. Inview of the foregoing, it is anticipated that the tendency to adopt PRMLat an order proportionately high to higher densities will continue.

Adopting high-order PRML suitable to intersymbol interference inresponse to enhancement of the recording line density can be describedas a method for recognizing a reproduced waveform using a waveformpattern in a longer period in order to identify a reproduced signal frominfluences of intersymbol interference by increasing the identificationresolution of the amplitude level of the reproduced waveform. Forexample, PR 12221 ML is PRML at a higher order than PR 1221 ML. Hence,by selecting suitable PRML according to a transmission path forinfluences of intersymbol interference caused by enhancement of therecording line density, it is possible to ensure the reproductionperformance.

An optical disc recording and reproducing device has to generate areproduction clock signal in synchronization with a reproduced signalduring the reproduction and to decode the reproduced signal into abinary digital signal in synchronization with the reproduction clocksignal. Generally, information about a reproduction clock signal iscontained at the edge of a recording mark. The optical disc recordingand reproducing device therefore generates the reproduction clock signalby detecting phase information about leading or lagging of the edge.However, in a case where a signal recorded, for example, at a recodingline density exceeding the limit of optical resolution is to bereproduced, there is a case where a reproduction clock signal cannot begenerated because the optical disc recording and reproducing devicefails to detect the phase information contained at the edgeappropriately due to influences of intersymbol interference.

SUMMARY OF THE INVENTION

The invention was devised to solve the problems discussed above and hasan object to provide a phase error detecting device, a waveform shapingdevice, and an optical disc device each capable of generating areproduction clock signal in a stable manner.

A phase error detecting device according to an aspect of the inventionincludes: a waveform shaping portion that receives a digital reproducedsignal generated from an analog reproduced signal reproduced from aninformation recording medium and shapes a waveform of the digitalreproduced signal; a maximum likelihood decoding portion that appliesmaximum likelihood decoding to the digital reproduced signal in thewaveform shaped by the waveform shaping portion and generates abinarized signal indicating a result of the maximum likelihood decoding;a phase detection portion that detects a phase error on the basis of thedigital reproduced signal in the waveform shaped by the waveform shapingportion and the binarized signal generated by the maximum likelihooddecoding portion; and a synchronization detection portion that generatesa reproduction clock signal using the phase error detected by the phasedetection portion and brings the digital reproduced signal insynchronization with the reproduction clock signal that has beengenerated. The phase detection portion extracts, during the maximumlikelihood decoding, the phase error using state transition patternshaving only a single zero cross point among differential metrics at aplurality of merging points at which a set of paths branched from agiven state merges.

According to this configuration, a digital reproduced signal generatedfrom an analog reproduced signal reproduced from an informationrecording medium is received and the waveform of the digital reproducedsignal is shaped by the waveform shaping portion. The digital reproducedsignal in the shaped waveform is then subjected to maximum likelihooddecoding by the maximum likelihood decoding portion. A binarized signalindicating the result of the maximum likelihood decoding is thusgenerated by the maximum likelihood decoding portion. Thereafter, aphase error is detected by the phase detection portion on the basis ofthe digital reproduced signal in the shaped waveform and the binarizedsignal that has been generated. In this instance, the phase error isextracted during the maximum likelihood decoding using the statetransition patterns having only a single zero cross point among thedifferential metrics at a plurality of merging points at which a set ofpaths branched from a given state merges. A reproduction clock signal isthen generated using the phase error that has been detected and thedigital reproduced signal is brought into synchronization with thereproduction clock signal that has been generated by the synchronizationdetection portion.

Accordingly, the phase error is extracted during the maximum likelihooddecoding using the state transition patterns having only a single zerocross point among the differential metrics at a plurality of mergingpoints at which a set of paths branched from a given state merges.Accuracy in detecting the phase error can be therefore enhanced, whichmakes it possible to generate a reproduction clock signal in a stablemanner.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a state transition diagram indicating a state transition ruledetermined from an RLL (1, 7) recoding code and equalization PR 12221ML;

FIG. 2 is a trellis diagram when the state transition diagram shown inFIG. 1 is developed with respect to a time axis;

FIG. 3 is a view showing the relation between a sample time and areproduction level (signal level) in transition paths set forth in Table1 below;

FIG. 4 is a view showing the relation between the sample time and thereproduction level (signal level) in transition paths set forth in Table2 below;

FIG. 5 is a view showing the relation between the sample time and thereproduction level (signal level) in transition paths set forth in Table3 below;

FIG. 6 is a block diagram showing the configuration of an optical discdevice according to a first embodiment of the invention;

FIG. 7 is a block diagram showing the configuration of a waveformshaping portion that uses an LMS algorithm;

FIG. 8 is a view showing an example of a variance (frequencydistribution) of a phase error found in accordance with Equation (2)below;

FIG. 9 is a view showing examples of an input waveform when the phasedisplaces significantly, an ideal waveform of a path A, and an idealwaveform of a path B;

FIG. 10 is a block diagram showing the configuration of a waveformshaping portion that uses a frequency sampling algorithm;

FIG. 11 is a view showing gain characteristic target values atrespective frequencies of a 9-tap FIR filter (digital equalizer);

FIG. 12 is a view showing tap coefficients of a digital equalizer foundin accordance with Equation (5) below using the frequency samplingalgorithm on the basis of the gain characteristic target values;

FIG. 13 is a view showing the frequency characteristic of the digitalequalizer calculated from the tap coefficients;

FIG. 14A is a view used to describe amplitude detection for calculatingcoefficients using the frequency sampling algorithm;

FIG. 14B is a view showing the processing result by an LPF on respectiveamplitude levels shown in FIG. 14A;

FIG. 15 is a block diagram showing the configuration of an optical discdevice according to a second embodiment of the invention;

FIG. 16 is a view showing the relative relation between a mark sequencerecorded on a track of an optical disc and a light beam diameter; and

FIG. 17 is a view showing an OTF of a BD having a recording capacity of25 GB.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS OF THE INVENTION

Hereinafter, embodiments of the invention will be described withreference to the drawings. It should be appreciated that embodimentsbelow are concrete examples of the invention and are not intended tolimit the technical scope of the invention.

Before descriptions of the embodiments, an edge shifting detectionmethod for a recording mark using PRML, which is the point of theinvention, will be described in the following.

A concrete optical disc device adopts PR 12221 ML as the signalprocessing in a reproduction system and uses codes, such as an RLL (RunLength Limited) (1, 7) code, as a recording code. Firstly, PR 12221 MLwill be briefly described using FIG. 1 and FIG. 2.

According to PR 12221 ML, the number of states in the decoding portionis limited to 10 due to the combination of RLL (1, 7). The number ofpaths of state transition in PR 12221 ML is 16 and the reproductionlevel includes nine levels.

FIG. 1 is a state transition diagram showing a state transition ruledetermined by the RLL (1, 7) recording code and PR 12221 ML. FIG. 1shows a state transition diagram generally used when PRML is described.Herein, 10 states are expressed as follows by denoting a state S (0, 0,0, 0) at a given time as S0, a state S (0, 0, 0, 1) as S1, a state S (0,0, 1, 1,) as S2, a state S (0, 1, 1, 1) as S3, a state S (1, 1, 1, 1) asS4, a state S (1, 1, 1, 0) as S5, a state S (1, 1, 0, 0) as S6, a stateS (1, 0, 0, 0) as S7, a state S (1, 0, 0, 1) as S8, and a state S (0, 1,1, 0) as S9. Herein, “0” or “1” inside the parentheses in FIG. 1represents a signal sequence on the time axis and specifies a state ofthe possibility of transition from one state to another in the followingtime. Also, a trellis diagram as shown in FIG. 2 is obtained bydeveloping the state transition diagram of FIG. 1 with respect to thetime axis.

FIG. 2 is a trellis diagram obtained by developing the state transitiondiagram shown in FIG. 1 with respect to the time axis. In the statetransition of PR 12221 ML as shown in FIG. 2, there are a myriad ofstate transition patterns (combinations of states) that can take twostate transitions when a predetermined state at a given time transitionsto another predetermined state at a different time. When attention ispaid to particularly error-prone state transition patterns within alimited time range, the state transition patters are those as set forthas in Tables 1, 2, and 3 below in the case of PR 12221 ML.

TABLE 1 STATE RECORDING CODE TRANSITION (b_(k−i), . . . , b_(k)) k − 9 k− 8 k − 7 k − 6 k − 5 k − 4 k − 3 S0_(k−5) → S6_(k) (0, 0, 0, 0, 1, 1,1, 0, 0) S0 S1 S2 (0, 0, 0, 0, 0, 1, 1, 0, 0) S0 S0 S1 S0_(k−5) → S5_(k)(0, 0, 0, 0, 1, 1, 1, 1, 0) S0 S1 S2 (0, 0, 0, 0, 0, 1, 1, 1, 0) S0 S0S1 S0_(k−5) → S4_(k) (0, 0, 0, 0, 1, 1, 1, 1, 1) S0 S1 S2 (0, 0, 0, 0,0, 1, 1, 1, 1) S0 S0 S1 S2_(k−5) → S0_(k) (0, 0, 1, 1, 1, 0, 0, 0, 0) S2S3 S5 (0, 0, 1, 1, 0, 0, 0, 0, 0) S2 S9 S6 S2_(k−5) → S1_(k) (0, 0, 1,1, 1, 0, 0, 0, 1) S2 S3 S5 (0, 0, 1, 1, 0, 0, 0, 0, 1) S2 S9 S6 S2_(k−5)→ S2_(k) (0, 0, 1, 1, 1, 0, 0, 1, 1) S2 S3 S5 (0, 0, 1, 1, 0, 0, 0,1, 1) S2 S9 S6 S3_(k−5) → S0_(k) (0, 1, 1, 1, 1, 0, 0, 0, 0) S3 S4 S5(0, 1, 1, 1, 0, 0, 0, 0, 0) S3 S5 S6 S3_(k−5) → S1_(k) (0, 1, 1, 1, 1,0, 0, 0, 1) S3 S4 S5 (0, 1, 1, 1, 0, 0, 0, 0, 1) S3 S5 S6 S3_(k−5) →S2_(k) (0, 1, 1, 1, 1, 0, 0, 1, 1) S3 S4 S5 (0, 1, 1, 1, 0, 0, 0, 1, 1)S3 S5 S6 S7_(k−5) → S6_(k) (1, 0, 0, 0, 1, 1, 1, 0, 0) S7 S1 S2 (1, 0,0, 0, 0, 1, 1, 0, 0) S7 S0 S1 S7_(k−5) → S5_(k) (1, 0, 0, 0, 1, 1, 1, 1,0) S7 S1 S2 (1, 0, 0, 0, 0, 1, 1, 1, 0) S7 S0 S1 S7_(k−5) → S4_(k) (1,0, 0, 0, 1, 1, 1, 1, 1) S7 S1 S2 (1, 0, 0, 0, 0, 1, 1, 1, 1) S7 S0 S1S6_(k−5) → S6_(k) (1, 1, 0, 0, 1, 1, 1, 0, 0) S6 S8 S2 (1, 1, 0, 0, 0,1, 1, 0, 0) S6 S7 S1 S6_(k−5) → S5_(k) (1, 1, 0, 0, 1, 1, 1, 1, 0) S6 S8S2 (1, 1, 0, 0, 0, 1, 1, 1, 0) S6 S7 S1 S6_(k−5) → S4_(k) (1, 1, 0, 0,1, 1, 1, 1, 1) S6 S8 S2 (1, 1, 0, 0, 0, 1, 1, 1, 1) S6 S7 S1 S4_(k−5) →S0_(k) (1, 1, 1, 1, 1, 0, 0, 0, 0) S4 S4 S5 (1, 1, 1, 1, 0, 0, 0, 0, 0)S4 S5 S6 S4_(k−5) → S1_(k) (1, 1, 1, 1, 1, 0, 0, 0, 1) S4 S4 S5 (1, 1,1, 1, 0, 0, 0, 0, 1) S4 S5 S6 S4_(k−5) → S2_(k) (1, 1, 1, 1, 1, 0, 0,1, 1) S4 S4 S5 (1, 1, 1, 1, 0, 0, 0, 1, 1) S4 S5 S6 STATE PREQUALIZATION EUCLIDEAN DISTANCE TRANSITION k − 2 k − 1 k IDEAL VALUEBETWEEN PATHS S0_(k−5) → S6_(k) S3 S5 S6 1 3 5 6 5 S2 S9 S6 0 1 3 4 5 14S0_(k−5) → S5_(k) S3 S4 S5 1 3 5 7 8 S2 S3 S5 0 1 3 5 7 14 S0_(k−5) →S4_(k) S3 S4 S4 1 3 5 7 8 S2 S3 S4 0 1 3 5 7 14 S2_(k−5) → S0_(k) S6 S7S0 5 6 5 3 1 S7 S0 S0 4 4 3 1 0 14 S2_(k−5) → S1_(k) S6 S7 S1 5 6 5 3 2S7 S0 S1 4 4 3 1 1 14 S2_(k−5) → S2_(k) S6 S8 S2 5 6 5 4 4 S7 S1 S2 4 43 2 3 14 S3_(k−5) → S0_(k) S6 S7 S0 7 7 5 3 1 S7 S0 S0 6 5 3 1 0 14S3_(k−5) → S1_(k) S6 S7 S1 7 7 5 3 2 S7 S0 S1 6 5 3 1 1 14 S3_(k−5) →S2_(k) S6 S8 S2 7 7 5 4 4 S7 S1 S2 6 5 3 2 3 14 S7_(k−5) → S6_(k) S3 S5S6 2 3 5 6 5 S2 S9 S6 1 1 3 4 4 14 S7_(k−5) → S5_(k) S3 S4 S5 2 3 5 7 7S2 S3 S5 1 1 3 5 6 14 S7_(k−5) → S4_(k) S3 S4 S4 2 3 5 7 8 S2 S3 S4 1 13 5 7 14 S6_(k−5) → S6_(k) S3 S5 S6 4 4 5 6 5 S2 S9 S6 3 2 3 4 4 14S6_(k−5) → S5_(k) S3 S4 S5 4 4 5 7 7 S2 S3 S5 3 2 3 5 6 14 S6_(k−5) →S4_(k) S3 S4 S4 4 4 5 7 8 S2 S3 S4 3 2 3 5 7 14 S4_(k−5) → S0_(k) S6 S7S0 8 7 5 3 1 S7 S0 S0 7 5 3 1 0 14 S4_(k−5) → S1_(k) S6 S7 S1 8 7 5 3 2S7 S0 S1 7 5 3 1 1 14 S4_(k−5) → S2_(k) S6 S8 S2 8 7 5 4 4 S7 S1 S2 7 53 2 3 14

TABLE 2 STATE RECORDING CODE TRANSITION (b_(k−i), . . . , b_(k)) k − 9 k− 8 k − 7 k − 6 k − 5 k − 4 k − 3 S0_(k−7) → S0_(k) (0, 0, 0, 0, 1, 1,0, 0, 0, 0, 0) S0 S1 S2 S9 S6 (0, 0, 0, 0, 0, 1, 1, 0, 0, 0, 0) S0 S0 S1S2 S9 S0_(k−7) → S1_(k) (0, 0, 0, 0, 1, 1, 0, 0, 0, 0, 1) S0 S1 S2 S9 S6(0, 0, 0, 0, 0, 1, 1, 0, 0, 0, 1) S0 S0 S1 S2 S9 S0_(k−7) → S2_(k) (0,0, 0, 0, 1, 1, 0, 0, 0, 1, 1) S0 S1 S2 S9 S6 (0, 0, 0, 0, 0, 1, 1, 0, 0,1, 1) S0 S0 S1 S2 S9 S2_(k−7) → S6_(k) (0, 0, 1, 1, 1, 0, 0, 1, 1, 0, 0)S2 S3 S5 S6 S8 (0, 0, 1, 1, 0, 0, 1, 1, 1, 0, 0) S2 S9 S6 S8 S2 S2_(k−7)→ S5_(k) (0, 0, 1, 1, 1, 0, 0, 1, 1, 1, 0) S2 S3 S5 S6 S8 (0, 0, 1, 1,0, 0, 1, 1, 1, 1, 0) S2 S9 S6 S8 S2 S2_(k−7) → S4_(k) (0, 0, 1, 1, 1, 0,0, 1, 1, 1, 1) S2 S3 S5 S6 S8 (0, 0, 1, 1, 0, 0, 1, 1, 1, 1, 1) S2 S9 S6S8 S2 S3_(k−7) → S6_(k) (0, 1, 1, 1, 1, 0, 0, 1, 1, 0, 0) S3 S4 S5 S6 S8(0, 1, 1, 1, 0, 0, 1, 1, 1, 0, 0) S3 S5 S6 S8 S2 S3_(k−7) → S5_(k) (0,1, 1, 1, 1, 0, 0, 1, 1, 1, 0) S3 S4 S5 S6 S8 (0, 1, 1, 1, 0, 0, 1, 1, 1,1, 0) S3 S5 S6 S8 S2 S3_(k−7) → S4_(k) (0, 1, 1, 1, 1, 0, 0, 1, 1, 1, 1)S3 S4 S5 S6 S8 (0, 1, 1, 1, 0, 0, 1, 1, 1, 1, 1) S3 S5 S6 S8 S2 S7_(k−7)→ S0_(k) (1, 0, 0, 0, 1, 1, 0, 0, 0, 0, 0) S7 S1 S2 S9 S6 (1, 0, 0, 0,0, 1, 1, 0, 0, 0, 0) S7 S0 S1 S2 S9 S7_(k−7) → S1_(k) (1, 0, 0, 0, 1, 1,0, 0, 0, 0, 1) S7 S1 S2 S9 S6 (1, 0, 0, 0, 0, 1, 1, 0, 0, 0, 1) S7 S0 S1S2 S9 S7_(k−7) → S2_(k) (1, 0, 0, 0, 1, 1, 0, 0, 0, 1, 1) S7 S1 S2 S9 S6(1, 0, 0, 0, 0, 1, 1, 0, 0, 1, 1) S7 S0 S1 S2 S9 S6_(k−7) → S0_(k) (1,1, 0, 0, 1, 1, 0, 0, 0, 0, 0) S6 S8 S2 S9 S6 (1, 1, 0, 0, 0, 1, 1, 0, 0,0, 0) S6 S7 S1 S2 S9 S6_(k−7) → S1_(k) (1, 1, 0, 0, 1, 1, 0, 0, 0, 0, 1)S6 S8 S2 S9 S6 (1, 1, 0, 0, 0, 1, 1, 0, 0, 0, 1) S6 S7 S1 S2 S9 S6_(k−7)→ S2_(k) (1, 1, 0, 0, 1, 1, 0, 0, 0, 1, 1) S6 S8 S2 S9 S6 (1, 1, 0, 0,0, 1, 1, 0, 0, 1, 1) S6 S7 S1 S2 S9 S4_(k−7) → S6_(k) (1, 1, 1, 1, 1, 0,0, 1, 1, 0, 0) S4 S4 S5 S6 S8 (1, 1, 1, 1, 0, 0, 1, 1, 1, 0, 0) S4 S5 S6S8 S2 S4_(k−7) → S5_(k) (1, 1, 1, 1, 1, 0, 0, 1, 1, 1, 0) S4 S4 S5 S6 S8(1, 1, 1, 1, 0, 0, 1, 1, 1, 1, 0) S4 S5 S6 S8 S2 S4_(k−7) → S4_(k) (1,1, 1, 1, 1, 0, 0, 1, 1, 1, 1) S4 S4 S5 S6 S8 (1, 1, 1, 1, 0, 0, 1, 1, 1,1, 1) S4 S5 S6 S8 S2 STATE PR EQUALIZATION EUCLIDEAN DISTANCE TRANSITIONk − 2 k − 1 k IDEAL VALUE BETWEEN PATHS S0_(k−7) → S0_(k) S7 S0 S0 1 3 44 3 1 0 S6 S7 S0 0 1 3 4 4 3 1 12 S0_(k−7) → S1_(k) S7 S0 S1 1 3 4 4 3 11 S6 S7 S1 0 1 3 4 4 3 2 12 S0_(k−7) → S2_(k) S7 S1 S2 1 3 4 4 3 2 3 S6S8 S2 0 1 3 4 4 4 4 12 S2_(k−7) → S6_(k) S2 S9 S6 5 6 5 4 4 4 4 S3 S5 S64 4 4 4 5 6 5 12 S2_(k−7) → S5_(k) S2 S3 S5 5 6 5 4 4 5 6 S3 S4 S5 4 4 44 5 7 7 12 S2_(k−7) → S4_(k) S2 S3 S4 5 6 5 4 4 5 7 S3 S4 S4 4 4 4 4 5 78 12 S3_(k−7) → S6_(k) S2 S9 S6 7 7 5 4 4 4 4 S3 S5 S6 6 5 4 4 5 6 5 12S3_(k−7) → S5_(k) S2 S3 S5 7 7 5 4 4 5 6 S3 S4 S5 6 5 4 4 5 7 7 12S3_(k−7) → S4_(k) S2 S3 S4 7 7 5 4 4 5 7 S3 S4 S4 6 5 4 4 5 7 8 12S7_(k−7) → S0_(k) S7 S0 S0 2 3 4 4 3 1 0 S6 S7 S0 1 1 3 4 4 3 1 12S7_(k−7) → S1_(k) S7 S0 S1 2 3 4 4 3 1 1 S6 S7 S1 1 1 3 4 4 3 2 12S7_(k−7) → S2_(k) S7 S1 S2 2 3 4 4 3 2 3 S6 S8 S2 1 1 3 4 4 4 4 12S6_(k−7) → S0_(k) S7 S0 S0 4 4 4 4 3 1 0 S6 S7 S0 3 2 3 4 4 3 1 12S6_(k−7) → S1_(k) S7 S0 S1 4 4 4 4 3 1 1 S6 S7 S1 3 2 3 4 4 3 2 12S6_(k−7) → S2_(k) S7 S1 S2 4 4 4 4 3 2 3 S6 S8 S2 3 2 3 4 4 4 4 12S4_(k−7) → S6_(k) S2 S9 S6 8 7 5 4 4 4 4 S3 S5 S6 7 5 4 4 5 6 5 12S4_(k−7) → S5_(k) S2 S3 S5 8 7 5 4 4 5 6 S3 S4 S5 7 5 4 4 5 7 7 12S4_(k−7) → S4_(k) S2 S3 S4 8 7 5 4 4 5 7 S3 S4 S4 7 5 4 4 5 7 8 12

TABLE 3 STATE RECORDING CODE TRANSITION (b_(k−i), . . . , b_(k)) k − 9 k− 8 k − 7 k − 6 k − 5 k − 4 k − 3 k − 2 S0_(k−9) → S6_(k) (0, 0, 0, 0,1, 1, 0, 0, 1, 1, 1, 0, 0) S0 S1 S2 S9 S6 S8 S2 S3 (0, 0, 0, 0, 0, 1, 1,0, 0, 1, 1, 0, 0) S0 S0 S1 S2 S9 S6 S8 S2 S0_(k−9) → S5_(k) (0, 0, 0, 0,1, 1, 0, 0, 1, 1, 1, 0, 1) S0 S1 S2 S9 S6 S8 S2 S3 (0, 0, 0, 0, 0, 1, 1,0, 0, 1, 1, 0, 1) S0 S0 S1 S2 S9 S6 S8 S2 S0_(k−9) → S4_(k) (0, 0, 0, 0,1, 1, 0, 0, 1, 1, 1, 1, 1) S0 S1 S2 S9 S6 S8 S2 S3 (0, 0, 0, 0, 0, 1, 1,0, 0, 1, 1, 1, 1) S0 S0 S1 S2 S9 S6 S8 S2 S2_(k−7) → S0_(k) (0, 0, 1, 1,1, 0, 0, 1, 1, 0, 0, 0, 0) S2 S3 S5 S6 S8 S2 S9 S6 (0, 0, 1, 1, 0, 0, 1,1, 0, 0, 0, 0, 0) S2 S9 S6 S8 S2 S9 S6 S7 S2_(k−7) → S1_(k) (0, 0, 1, 1,1, 0, 0, 1, 1, 0, 0, 0, 1) S2 S3 S5 S6 S8 S2 S9 S6 (0, 0, 1, 1, 0, 0, 1,1, 0, 0, 0, 0, 1) S2 S9 S6 S8 S2 S9 S6 S7 S2_(k−7) → S2_(k) (0, 0, 1, 1,1, 0, 0, 1, 1, 0, 0, 1, 1) S2 S3 S5 S6 S8 S2 S9 S6 (0, 0, 1, 1, 0, 0, 1,1, 0, 0, 0, 1, 1) S2 S9 S6 S8 S2 S9 S6 S7 S3_(k−5) → S0_(k) (0, 1, 1, 1,1, 0, 0, 1, 1, 0, 0, 0, 0) S3 S4 S5 S6 S8 S2 S9 S6 (0, 1, 1, 1, 0, 0, 1,1, 0, 0, 0, 0, 0) S3 S5 S6 S8 S2 S9 S6 S7 S3_(k−5) → S1_(k) (0, 1, 1, 1,1, 0, 0, 1, 1, 0, 0, 0, 1) S3 S4 S5 S6 S8 S2 S9 S6 (0, 1, 1, 1, 0, 0, 1,1, 0, 0, 0, 0, 1) S3 S5 S6 S8 S2 S9 S6 S7 S3_(k−5) → S2_(k) (0, 1, 1, 1,1, 0, 0, 1, 1, 0, 0, 1, 1) S3 S4 S5 S6 S8 S2 S9 S6 (0, 1, 1, 1, 0, 0, 1,1, 0, 0, 0, 1, 1) S3 S5 S6 S8 S2 S9 S6 S7 S3_(k−5) → S2_(k) (1, 0, 0, 0,1, 1, 0, 0, 1, 1, 1, 0, 0) S7 S1 S2 S9 S6 S8 S2 S3 (1, 0, 0, 0, 0, 1, 1,0, 0, 1, 1, 0, 0) S7 S0 S1 S2 S9 S6 S8 S2 S3_(k−5) → S2_(k) (1, 0, 0, 0,1, 1, 0, 0, 1, 1, 1, 1, 0) S7 S1 S2 S9 S6 S8 S2 S3 (1, 0, 0, 0, 0, 1, 1,0, 0, 1, 1, 1, 0) S7 S0 S1 S2 S9 S6 S8 S2 S3_(k−5) → S2_(k) (1, 0, 0, 0,1, 1, 0, 0, 1, 1, 1, 1, 1) S7 S1 S2 S9 S6 S8 S2 S3 (1, 0, 0, 0, 0, 1, 1,0, 0, 1, 1, 1, 1) S7 S0 S1 S2 S9 S6 S8 S2 S6_(k−5) → S6_(k) (1, 1, 0, 0,1, 1, 0, 0, 1, 1, 1, 0, 0) S6 S8 S2 S9 S6 S8 S2 S3 (1, 1, 0, 0, 0, 1, 1,0, 0, 1, 1, 0, 0) S6 S7 S1 S2 S9 S6 S8 S2 S6_(k−5) → S5_(k) (1, 1, 0, 0,1, 1, 0, 0, 1, 1, 1, 1, 0) S6 S8 S2 S9 S6 S8 S2 S3 (1, 1, 0, 0, 0, 1, 1,0, 0, 1, 1, 1, 0 S6 S7 S1 S2 S9 S6 S8 S2 S6_(k−5) → S4_(k) (1, 1, 0, 0,1, 1, 0, 0, 1, 1, 1, 1, 1) S6 S8 S2 S9 S6 S8 S2 S3 (1, 1, 0, 0, 0, 1, 1,0, 0, 1, 1, 1, 1) S6 S7 S1 S2 S9 S6 S8 S2 S4_(k−5) → S0_(k) (1, 1, 1, 1,1, 0, 0, 1, 1, 0, 0, 0, 0) S4 S4 S5 S6 S8 S2 S9 S6 (1, 1, 1, 1, 0, 0, 1,1, 0, 0, 0, 0, 0) S4 S5 S6 S8 S2 S9 S6 S7 S4_(k−5) → S1_(k) (1, 1, 1, 1,1, 0, 0, 1, 1, 0, 0, 0, 1) S4 S4 S5 S6 S8 S2 S9 S6 (1, 1, 1, 1, 0, 0, 1,1, 0, 0, 0, 0, 1) S4 S5 S6 S8 S2 S9 S6 S7 S4_(k−5) → S2_(k) (1, 1, 1, 1,1, 0, 0, 1, 1, 0, 0, 1, 1) S4 S4 S5 S6 S8 S2 S9 S6 (1, 1, 1, 1, 0, 0, 1,1, 0, 0, 0, 1, 1) S4 S5 S6 S8 S2 S9 S6 S7 STATE PR EQUALIZATIONEUCLIDEAN DISTANCE TRANSITION k − 1 k IDEAL VALUE BETWEEN PATHS S0_(k−9)→ S6_(k) S5 S6 1 3 4 4 4 4 5 6 5 S9 S6 0 1 3 4 4 4 4 4 4 12 S0_(k−9) →S5_(k) S4 S5 1 3 4 4 4 4 5 7 7 S3 S5 0 1 3 4 4 4 4 5 6 12 S0_(k−9) →S4_(k) S4 S4 1 3 4 4 4 4 5 7 8 S3 S4 0 1 3 4 4 4 4 5 7 12 S2_(k−7) →S0_(k) S7 S0 5 6 5 4 4 4 4 3 1 S0 S0 4 4 4 4 4 4 3 1 0 12 S2_(k−7) →S1_(k) S7 S1 5 6 5 4 4 4 4 3 2 S0 S1 4 4 4 4 4 4 3 1 1 12 S2_(k−7) →S2_(k) S8 S2 5 6 5 4 4 4 4 4 4 S1 S2 4 4 4 4 4 4 3 2 3 12 S3_(k−5) →S0_(k) S7 S0 7 7 5 4 4 4 4 3 1 S0 S0 6 5 4 4 4 4 3 1 0 12 S3_(k−5) →S1_(k) S7 S1 7 7 5 4 4 4 4 3 2 S0 S1 6 5 4 4 4 4 3 1 1 12 S3_(k−5) →S2_(k) S8 S2 7 7 5 4 4 4 4 4 4 S1 S2 6 5 4 4 4 4 3 2 3 12 S3_(k−5) →S2_(k) S5 S6 2 3 4 4 4 4 5 6 5 S9 S6 1 1 3 4 4 4 4 4 4 12 S3_(k−5) →S2_(k) S4 S5 2 3 4 4 4 4 5 7 7 S3 S5 1 1 3 4 4 4 4 5 6 12 S3_(k−5) →S2_(k) S4 S4 2 3 4 4 4 4 5 7 8 S3 S4 1 1 3 4 4 4 4 5 7 12 S6_(k−5) →S6_(k) S5 S6 4 4 4 4 4 4 5 6 5 S9 S6 3 2 3 4 4 4 4 4 4 12 S6_(k−5) →S5_(k) S4 S5 4 4 4 4 4 4 5 7 7 S3 S5 3 2 3 4 4 4 4 5 6 12 S6_(k−5) →S4_(k) S4 S4 4 4 4 4 4 4 5 7 8 S3 S4 3 2 3 4 4 4 4 5 7 12 S4_(k−5) →S0_(k) S7 S0 8 7 5 4 4 4 4 3 1 S0 S0 7 5 4 4 4 4 3 1 0 12 S4_(k−5) →S1_(k) S7 S1 8 7 5 4 4 4 4 3 2 S0 S1 7 5 4 4 4 4 3 1 1 12 S4_(k−5) →S2_(k) S8 S2 8 7 5 4 4 4 4 4 4 S1 S2 7 5 4 4 4 4 3 2 3 12

Set forth in each table above are state transitions indicating a tracefrom a start state to a merged state, two recording sequences that arelikely when the state undergoes the specified transitions, two idealreproduced waveforms that are likely when the state undergoes thespecified transitions, and a Euclidean distance between two idealreproduced waveforms.

The Euclidean distance indicates a sum of squares of a differencebetween two ideal reproduced waveforms. When the likelihood of tworeproduced waveforms is determined, one is distinguished from the othermore readily when the value of the Euclidean distance is large and apossibility of erroneous determination is reduced. Meanwhile, when thevalue of the Euclidean distance is small, it becomes difficult todistinguish between two likely waveforms and a possibility of erroneousdetermination becomes higher. In other words, a state transition patternhaving a large Euclidean distance can be said as a less error-pronestate transition pattern and a state transition pattern having a smallEuclidean distance can be said as an error-prone state transitionpattern.

Table 1 above shows state transition patterns that can take two statetransitions and it shows state transition patterns in a case where aEuclidean distance is 14. There are 18 state transition patterns in acase where a Euclidean distance is 14. The state transition patterns setforth in Table 1 above correspond to edge (switching between a mark anda space) portions of a waveform of an optical disc. In other words, thestate transition patterns set forth in Table 1 are patterns of 1-bitshift error of the edge.

By way of example, transition paths in a case where the state S0 (k−5)transitions to the state S6 (k) according to the state transition ruleshown in FIG. 2 will be described.

Herein, one transition path is a case where the recording sequence thathas transitioned to “0, 0, 0, 0, 1, 1, 1, 0, 0” is detected. Assume thata “0” in the reproduced data is a space portion and “1” is a markportion and this transition path is in a recording state. Then, therecording state is a combination of a space of a length of 4 T space ofmore, a 3 T mark, and a space of a length of 2 T space of more. Therelation between the sample time and the reproduction level (signallevel) in this transition path is shown in FIG. 3 as a path A waveform.

FIG. 3 is a view showing the relation between the sample time and thereproduction level (signal level) in the transition paths set forth inTable 1 above. In FIG. 3 through FIG. 5, the abscissa is used for thesample time per clock time of the recording sequence and the ordinate isused for the reproduction level. Although it has been described above,in the case of PR 12221 ML, the ideal reproduction levels are ninelevels from a level 0 to a level 8.

The other transition path is a case where the recording sequence thathas transitioned to “0, 0, 0, 0, 0, 1, 1, 0, 0” is detected. Assume that“0” in the reproduced data is a space portion and “1” is a mark portionand this transition path is in a recording state. Then, the recordingstate is a combination of a space of a length of 5 T space of more, a 2T mark, and a space of a length of 2 T space or more. The relationbetween the sample time and the reproduction level (signal level) inthis transition path is shown in FIG. 3 as a path B waveform. The statetransition patterns with a Euclidean distance of 14 set forth in table 1above are characterized in that they are patterns each necessarilycontaining one piece of edge information (zero cross point).

Table 2 above shows state transition patterns that can take two statetransitions as those in Table 1 above and it shows state transitionpatterns in a case where a Euclidean distance is 12. There are 18 statetransition patterns in a case where a Euclidean distance is 12. Thestate transition patterns set forth in Table 2 above are shift errors ofa 2 T mark or a 2 T space and they are patterns of a 2-bit shift error.

By way of example, transition paths in a case where the state S0 (k−7)transitions to the state S0 (k) according to the state transition ruleshown in FIG. 2 will be described.

It should be noted, however, that FIG. 2 only shows states up to thetime k−5. Herein, one transition path is a case where the recordingsequence that has transitioned to “0, 0, 0, 0, 1, 1, 0, 0, 0, 0, 0” isdetected. Assume that “0” in the reproduced data is a space portion and“1” is a mark portion and this transition path is in a recording state.Then, the recording state is a combination of a space of a length of 4 Tspace of more, a 2 T mark, and a space of a length of 5 T space of more.The relation between the sample time and the reproduction level (signallevel) in this transition path is shown in FIG. 4 as a path A waveform.FIG. 4 is a view showing the relation between the sample time and thereproduction level (signal level) in the transition paths set forth inTable 2 above.

The other transition path is a case where the recording sequence thathas transitioned to “0, 0, 0, 0, 0, 1, 1, 0, 0, 0, 0” is detected.Assume that “0” in the reproduced data is a space portion and “1” is amark portion and this transition path is in a recording state. Then, therecording state is a combination of a space of a length of 5 T space ofmore, a 2 T mark, and a space of a length of 4 T space of more. Therelation between the sample time and the reproduction level (signallevel) in this transition path is shown in FIG. 4 as a path B waveform.The state transition patterns with a Euclidean distance of 12 set forthin Table 2 above are characterized in that they are patterns eachnecessarily containing two pieces of edge information about the risingand the falling of 2 T.

Table 3 above shows state transition patterns that can take two statetransitions as those in Table 1 and Table 2 above and it shows the statetransition patterns in a case where a Euclidean distance is 12. Thereare 18 state transition patterns in a case where a Euclidean distance is12. The state transition patterns set forth in Table 3 above are pointsat which a 2 T mark and a 2 T space are continuous and they are patternsof a 3-bit shift error.

By way of example, transition paths in a case where the state S0 (k−9)transitions to the state S6 (k) according to the state transition ruleshown in FIG. 2 will be described.

It should be noted, however, that FIG. 2 only shows the states up to thetime k−5. Herein, one transition path is a case where the recordingsequence that has transitioned to “0, 0, 0, 0, 1, 1, 0, 0, 1, 1, 1, 0,0” is detected. Assume that “0” in the reproduced data is a spaceportion and “1” is a mark portion and this transition path is in arecording state. Then, the recording state is a combination a space of alength of a 4 T space of more, a 2 T mark, a 2 T space, a 3 T mark, anda space of a length of a 2 T space of more. The relation between thesample time and the reproduction level (signal level) in this transitionpath is shown in FIG. 5 as a path A waveform. FIG. 5 is a view showingthe relation between the sample time and the reproduction level (signallevel) in the transition paths set forth in Table 3 above.

The other transition path is a case where the recording sequence thathas transitioned to “0, 0, 0, 0, 0, 1, 1, 0, 0, 1, 1, 0, 0” is detected.Assume that “0” in the reproduced data is a space portion and “1” is amark portion and this transition path is in a recording state. Then, therecording state is a combination of a space of a length of a 5 T spaceof more, a 2 T mark, a 2 T space, a 2 T mark, and a space of a length ofa 2 T space of more. The relation between the sample time and thereproduction level (signal level) in this transition path is shown inFIG. 5 as a path B waveform. The state transition patterns with aEuclidean distance of 12 set forth in Table 3 above are characterized inthat they are patterns each containing at least three pieces of edgeinformation.

In a case where a synchronous signal of a reproduced signal is generatedusing the displacement information of the leading edge or the trailingedge of a mark of a signal to be reproduced, it is necessary that thephase displacement direction can be detected separately at the leadingedge portion and the trailing edge portion of each mark length. In acase where PR 12221 ML is used, it becomes possible to detect a phaseerror for synchronization with a reproduced signal using statetransition patterns with a Euclidean distance of 14 described withreference to Table 1 above.

First Embodiment

Initially, an optical disc device according to a first embodiment of theinvention will be described. FIG. 6 is a block diagram showing theconfiguration of the optical disc device according to the firstembodiment of the invention.

An information recording medium 101 is an information recording mediumin and from which information is optically recorded and reproduced, andit is, for example, an optical disc.

The optical disc device 100 shown in FIG. 6 includes an optical headportion 102, a pre-amplifier portion 103, an AGC (Automatic GainController)/OFFSET portion 104, a waveform equalization portion 105, anA/D conversion portion 106, an AGC/OFFSET control portion 107, awaveform shaping portion 108, a maximum likelihood decoding portion 111,a phase detection portion 112, and a synchronization detection portion115. Further, the waveform shaping portion 108 includes a PR (PartialResponse) equalization portion 109 and an adaptive coefficient updateportion 110. The phase detection portion 112 includes a particularpattern detection portion 113 and a differential metric detectionportion 114. The synchronization detection portion 115 includes a clockcontrol portion 116 and a synchronization entrainment control portion117.

The optical head portion 102 generates an analog reproduced signalexpressing information read out from the information recording medium101. The pre-amplifier portion 103 amplifies the analog reproducedsignal generated in the optical head portion 102 by a predetermined gainand outputs the resulting signal to the AGC/OFFSET portion 104.

The AGC/OFFSET portion 104 amplifies or attenuates the analog reproducedsignal from the pre-amplifier portion 103 according to a control signalfrom the AGC/OFFSET control portion 107 so that the analog reproducedsignal has a predetermined amplitude and outputs the resulting signal tothe waveform equalization portion 105.

The waveform equalization portion 105 has a filter characteristic toamplify a high frequency of the analog reproduced signal and therebyamplifies a high frequency portion of the reproduced waveform. It thenoutputs the result to the A/D conversion portion 106.

The AGC/OFFSET control portion 107 outputs a control signal to theAGC/OFFSET portion 104 so that the level of a digital reproduced signaloutputted from the A/D conversion portion 106 reaches a pre-set targetlevel.

The A/D conversion portion 106 converts an analog reproduced signal to adigital reproduced signal by sampling the analog reproduced signal insynchronization with a reproduction clock outputted from thesynchronization detection portion 115 and outputs the result to thewaveform shaping portion 108.

The waveform shaping portion 108 receives a digital reproduced signalgenerated from an analog reproduced signal reproduced from theinformation recording medium 111 and shapes the waveform of the digitalreproduced signal.

The PR equalization portion 109 is furnished with a capability of makingthe filter characteristic variable to the characteristics of varioustypes of PR. The PR equalization portion 109 applies filtering in orderto achieve the frequency characteristic set so that the frequencycharacteristic of the recoding and reproducing system will be thecharacteristic assumed by the maximum likelihood decoding portion 111(for example, PR (1, 2, 2, 2, 1) equalization characteristic). It thenapplies PR equalization, which is a process to suppress high-frequencynoises and intentionally append intersymbol interference, to a digitalreproduced signal and outputs the resulting signal to the maximumlikelihood decoding portion 111. The PR equalization portion 109 isformed, for example, of an FIR (Finite Impulse Response) filter.

The adaptive coefficient update portion 110 changes and updatescoefficients of the FIR filter in the PR equalization portion 109according to the characteristic of a reproduced signal inputted thereinso that an output of the PR equalization portion 109 will have a desiredPR characteristic.

The maximum likelihood decoding portion 111 applies maximum likelihooddecoding to the digital reproduced signal in the waveform shaped by thewaveform shaping portion 108 and thereby generates a binarized signalthat indicates the result of the maximum likelihood decoding. Themaximum likelihood decoding portion 111 is, for example, a Viterbidecoder. It uses the maximum likelihood decoding method by which a mostlikely sequence is predicted according to the coding rule appendedintentionally in response to the type of the partial response (PR)characteristic to decode the reproduced signal subjected to PRequalization in the PR equalization portion 109, and thereby outputsbinarized data. The binarized data outputted from the maximum likelihooddecoding portion 111 is outputted to a circuit (not shown) in the latterstage as a decoded binarized signal. A predetermined process is appliedto this data and information recorded in the information recordingmedium 101 is reproduced.

The phase detection portion 112 detects a phase error on the basis thedigital reproduced signal in the waveform shaped by the waveform shapingportion 108 and the binarized signal generated by the maximum likelihooddecoding portion 111. Also, the phase detection portion 112 extracts,during the maximum likelihood decoding, a phase error using statetransition patterns having only a single zero cross point amongdifferential metrics at a plurality of merging points at which a set ofpaths branched from a given state will merge.

The particular pattern detection portion 113 in the phase detectionportion 112 detects particular state transition patterns from thebinarized data outputted from the maximum likelihood decoding portion111. As a concrete example, it detects state transition patterns of 36recording coding sequences (bk−i, . . . , and bk) set forth in Table 1above. It is configured in such a manner that the particular statetransition patterns to be detected can be adaptively selected. Theparticular pattern detection portion 113 is configured in such a mannerthat it is able to select state transition patterns used to calculate aphase error in response to circumstances in the synchronizationdetection portion 115, such as during frequency entrainment, duringphase entrainment, and at the occurrence of deterioration inreproduction quality.

The differential metric detection portion 114 in the phase detectionportion 112 calculates differential metrics only in the patternsdetected by the particular pattern detection portion 113 and furtherconverts the differential metrics thus calculated into phase errorinformation.

The synchronization detection portion 115 generates a reproduction clocksignal using the phase error detected by the phase detection portion 112and brings the digital reproduced signal into synchronization with thereproduction clock signal thus generated.

The clock control portion 116 in the synchronization detection portion115 converts an output of the phase detection portion 112 into thefrequency of the reproduction clock by allowing the phase errorinformation inputted therein from the phase detection portion 112 topass through a predetermined filter and outputs the convertedreproduction clock to the A/D conversion portion 106. The clock controlportion 116 may use a VCO (Voltage Control Oscillator) in order toconvert a phase error into the frequency of the reproduction clock.

The synchronization entrainment control portion 117 in thesynchronization detection portion 115 outputs signals that controlrespective blocks during synchronization entrainment. Thesynchronization entrainment control portion 117 controls operations ofthe AGC/OFFSET control portion 107, the adaptive coefficient updateportion 110, and the phase detection portion 112.

The detailed configuration of the waveform shaping portion 108 shown inFIG. 6 will now be described. FIG. 7 is a block diagram showing theconfiguration of a waveform shaping portion that uses an LMS algorithm.The first embodiment is configured to update filter coefficients usingthe LMS (Least Mean Square) algorithm. The setup equation ofequalization coefficients in the LMS algorithm is expressed as Equation(1) below. The LMS algorithm is disclosed in JP-A-2003-85764.

w(n(T+1))=w(nT)+A·e(nT)·x(nT)  (1)

where T=0, 1, 2, 3, and so on.

Herein, w(nT) is a current coefficient, w(n(T+1)) is an updatedcoefficient, A is a tap gain, e(nT) is an equalization error, x(nT) isan FIR filter input signal. Also, n is a parameter to select an updatecycle of the coefficient. In accordance with Equation (1) above, theequalization coefficients of the FIR (Finite Impulse Response) filterare updated.

In this embodiment, the filter coefficients are updated using the LMSalgorithm. It should be appreciated, however, that the algorithm is notparticularly limited to an update control algorithm of the filtercoefficients and other algorithms may be used as well.

The PR equalization portion 109 includes the FIR filter portion 201. Theadaptive coefficient update portion 110 includes an error signaldetection portion 202, a correlation detection portion 203, a loop gainsetting portion 204, and a coefficient computation portion 205. Into thewaveform shaping portion 108 are inputted a digital reproduced signaloutputted from the A/D conversion portion 106 and a PR equalizationteacher signal outputted from an unillustrated predetermined circuit.From the waveform shaping portion 108 is outputted a digital filtersignal after desired filter processing is applied thereon throughvarious operations according to a control signal from thesynchronization entrainment control portion 117.

The error signal detection portion 202 calculates the equalization error(e(nT)) in Equation (1) above. The error signal detection portion 202outputs a difference between the PR equalization teacher signal and anoutput of the FIR filter portion 201 (digital filter output signal)inputted therein as an error signal. The correlation detection portion203 calculates a product value (e (nT)×(nT)) of the equalization errorin Equation (1) above and a digital reproduced signal. The loop gainsetting portion 204 sets a loop gain of a coefficient update loop in theFIR filter portion 201 by multiplying an output of the correlationdetection portion 203 by the tap gain (A) in Equation (1) above.

The coefficient computation portion 205 updates coefficients of thefilter on the basis of an error signal from the loop gain settingportion 204 and outputs the updated coefficients to the FIR filterportion 201. The coefficient computation portion 205 calculates eachupdated filter coefficient (w(n(T+1))) by adding the current filtercoefficient (w(nT)) to an output of the loop gain setting portion 204.

The adaptive coefficient update portion 110 of this embodiment is formedwithin a PLL (Phase Locked Loop) loop for synchronization with areproduced signal and therefore has two points as follows.

A first point is a coefficient update control method. The coefficientcomputation portion 205 can have a capability of individuallycontrolling as many coefficients as tap coefficients (tap gains)provided to the FIR filter portion 201, but it is characterized in thatit controls the coefficients to be right-left symmetrical. Thecoefficient control under the restriction of right-left symmetryreferred to herein means the control under which the gain at apredetermined frequency alone can be changed without having to changethe phase relation between a digital reproduced signal inputted into theFIR filter portion 201 and a digital filter output signal outputted fromthe FIR filter portion 201.

Because the waveform shaping portion 108 is disposed within the PLLloop, this control is performed with the purpose not to correct thephase in the FIR filter portion 201 so that a phase error can bedetected appropriately in the phase detection portion 112 in the latterstage of the FIR filter portion 201. As an example of the coefficientcontrol that makes the tap coefficients right-left symmetrical, there isa method by which an average of the tap coefficients in the right andleft is calculated after a plurality of tap coefficients in the FIRfilter portion 201 are calculated and the result of calculation is usedin the FIR filter portion 201 as the tap coefficients. It should beappreciated, however, that this embodiment is not limited to thisexample of the coefficient control. The coefficient control to make thetap coefficients right-left symmetrical may be performed using othercalculation methods.

A second point is a detection method of an equalization error found inthe error signal detection portion 202. Regarding the detection of anequalization error, it is general that an output of the FIR filterportion 201 is converted into a binary form by a predetermined method togenerate an ideal waveform from the resulting binarized signal and adifference between the ideal waveform and an output waveform of the FIRfilter portion 201 is detected as an equalization error. On thecontrary, this embodiment is characterized in that an equalization erroris not detected from a signal waveform relating to 2 T or 3 T and from asignal waveform having an error as large as or larger than apredetermined error.

Initially, regarding the binarization of an output of the FIR filterportion 201, it is possible to enhance the binarization accuracy bydecoding the output using the maximum likelihood decoding, such asViterbi decoding. From the viewpoint that the error signal detectionportion 202 performs processing within the PLL loop, it is necessary toprovide the configuration that takes latency into account (to providethe configuration in which a delay is suppressed to the minimum).Accordingly, the binarization to generate an ideal waveform is performedusing a waveform after the FIR filtering, for example, by the leveldetermination processing. The binarization accuracy cannot be thereforeexpected. Further, as has been described above, in a case where a regionin which information is recorded at a higher recording line density isreproduced, much less is expected for binarization accuracy with thereproduced waveform.

In view of the foregoing, this embodiment makes it possible to stabilizean adaptive equalization loop not by detecting an equalization errorfrom a signal waveform relating to 2 T or 3 T and from a signal waveformhaving an error as large as or larger than a predetermined error. In acase where an equalization error is not detected from a signal waveformrelating to 2 T or 3 T, there may be produced a side-effect that thecoefficients cannot be controlled so as to have a predetermined PRcharacteristic. However, the phase information cannot be extracted, inparticular, from a reproduced waveform of a 2 T signal in which thefrequency of the shortest mark is in the vicinity of an OTF cutofffrequency or exceeds the OTF cutoff frequency. In addition, although itdepends on a recording line density, there is a case where it becomesdifficult to extract the phase information from a 3 T signal because theamplitude is extremely small.

Accordingly, by setting a pattern not to extract a phase error in thephase detection portion 112 in the latter stage, there is no need tomake the coefficients to have the predetermined PR characteristic athigh accuracy for a 2 T signal and a 3 T signal. In addition, becausethe frequency characteristics expressed by the FIR filter are limited bylimiting the number of taps, the coefficients are corrected to someextent. For example, the number of taps may be set to 5 to 13. Hence, itis not that the gain correction of a 2 T signal and a 3 T signal is nottaken into account at all.

It should be appreciated that the detection control of an equalizationerror in this embodiment is not limited to this example of detectioncontrol. For example, it may be the detection control under which anequalization error is not detected from the signal waveform relating toa 2 T signal alone.

Further, it is configured in such a manner that a pattern from which todetect a phase error can be selected adaptively in the phase detectionportion 112. Hence, this embodiment is not limited to the exampledescribed above.

An operation of the optical disc device configured as above will now bedescribed.

As is set forth in Table 4 below, the synchronization entrainmentcontrol portion 117 in the synchronization detection portion 115controls the AGC/OFFSET control portion 107, the adaptive coefficientupdate portion 110 in the waveform shaping portion 108, and theparticular pattern detection portion 113 in the phase detection portion112.

TABLE 4 ADAPTIVE AGC/OFFSET COEFFICIENT CONTROL UPDATE PARTICULARPATTERN CONTROL STATE PORTION 107 PORTION 110 DETECTION PORTION 113BEFORE TRACKING OFF OFF OFF CONTROL IS ENABLED BEFORE AGC/OFFSET ON OFFOFF CONVERGENCE IS ENABLED BEFORE FREQUENCY ON ON OFF (DETECTENTRAINMENT IS ENABLED SYNCHRONIZATION MARK) BEFORE PHASE ON ON TABLE 1ENTRAINMENTIS ENABLED (ALL PATTERNS) AFTER PHASE ON ON TABLE 1ENTRAINMENTIS ENABLED (DELETE 2T PATTERNS)

As examples of the control state, states are classified into statesbefore the tracking control is enabled, before the AGC/OFFSETconvergence is enabled, before the frequency entrainment is enabled,before the phase entrainment is enabled, and after the phase entrainmentis enabled, each of which will be described in detail below.

A reproduced signal having a predetermined amplitude is obtained fromthe preamplifier portion 103 by performing the focus control and thetracking control in such a manner that the optical head portion 102 iscontrolled to move to the information recording medium 101 at a regionin which information is recorded and the motor that rotates theinformation recording medium 101 is controlled to run so that theinformation recording medium 101 rotates at a predetermined rotationnumber.

When the tracking control becomes an ON state, the AGC/OFFSET controlportion 107 starts the control and outputs a control signal to theAGC/OFFSET portion 104 so that the level of a digital signal outputtedfrom the A/D conversion portion 106 reaches the pre-set target level.

The AGC/OFFSET control portion 107 outputs an AGC/OFFSET portion controlenable signal to the synchronization detection portion 115 when thelevel of a digital reproduced signal outputted from the A/D conversionportion 106 falls within a predetermined range with respect to thetarget level.

Until the control on the AGC/OFFSET portion 104 is enabled, thesynchronization entrainment control portion 117 in the synchronizationdetection portion 115 outputs control signals to inhibit the control onthe adaptive coefficient update portion 110 in the waveform shapingportion 108, the phase detection in the phase detection portion 112, andthe frequency control and the phase control in the synchronizationdetection portion 115.

When the control on the AGC/OFFSET portion 104 is enabled, thesynchronization entrainment control portion 117 starts the frequencycontrol first. Herein, detailed descriptions of the frequency controlare omitted and an operation thereof will be described briefly.Reference should be made to JP-A-2006-504217 (kohyo) for details.

When the frequency control is started, the adaptive coefficient updateportion 110 starts the FIR coefficient control and inputs the waveformafter the predetermined PR equalization into the maximum likelihooddecoding portion 111. The maximum likelihood decoding portion 111 thenconverts the signal inputted therein into a binary form. The binarizedsignal outputted from the maximum likelihood decoding portion 111 isinputted into the phase detection portion 112. The phase detectionportion 112 then detects a synchronization mark for detecting afrequency error. Although the synchronization mark is not shown in thedrawings, in the case of a BD (Blu-ray disc), the synchronization markis a 9 T9 T pattern and the data is a combined pattern from 2 T to 8 T.

The phase detection portion 112 detects a continuous pattern length thatwill be the longest in a period that is several times the interval inwhich the synchronization mark is inserted and outputs a pattern lengtherror (frequency error) between the pattern length thus detected and the9 T9 T pattern length to the synchronization detection portion 115. Theclock control portion 116 in the synchronization detection portion 115varies the oscillation frequency of the VCO according to the frequencyerror inputted therein from the phase detection portion 112. When thefrequency error is equal to or smaller the predetermined error, theclock control portion 116 determines that the frequency entrainment isenabled and starts the phase entrainment subsequently.

When the phase entrainment is started, the particular pattern detectionportion 113 in the phase detection portion 112 detects a total of 36state transition patterns set forth in Table 1 above from the binarizedsequences inputted therein from the maximum likelihood decoding portion111. Subsequently, the differential metric detection portion 114calculates a differential metric error of the state transition patternsdetected by the particular pattern detection portion 113, converts thedifferential metric error thus calculated into phase information, andoutputs the phase information to the synchronization detection portion115. The clock control portion 116 in the synchronization detectionportion 115 then varies the oscillation frequency of the VCO accordingto the phase information (error) inputted therein from the phasedetection portion 112. For example, the clock control portion 116determines that the phase entrainment is enabled (completed) when thesynchronization mark is detected a predetermined number of times insuccession.

When the phase entrainment is completed, the particular patterndetection portion 113 in the phase detection portion 112 detects thestate transition patterns other than a total of 10 state transitionpatterns relating to the 2 T pattern among a total of 36 statetransition patterns set forth in Table 1 above from the binarizedsequences inputted therein from the maximum likelihood decoding portion111. Subsequently, the differential metric detection portion 114calculates a differential metric error among the detected statetransition patterns, converts the differential metric error thuscalculated into phase information, and outputs the phase information tothe synchronization detection portion 115. The clock control portion 116in the synchronization detection portion 115 then controls theoscillation frequency of the VCO according to the phase information(error) inputted therein from the phase detection portion 112.

Herein, a total of 10 state transition patterns relating to the 2 Tpattern are (0, 0, 0, 0, 0, 1, 1, 0, 0), (0, 0, 1, 1, 0, 0, 0, 0, 0),(0, 0, 1, 1, 0, 0, 0, 0, 1), (0, 0, 1, 1, 0, 0, 0, 1, 1), (0, 1, 1, 1,1, 0, 0, 1, 1,), (1, 0, 0, 0, 0, 1, 1, 0, 0), (1, 1, 0, 0, 1, 1, 1, 0,0), (1, 1, 0, 0, 1, 1, 1, 1, 0), (1, 1, 0, 0, 1, 1, 1, 1, 1), and (1, 1,1, 1, 1, 0, 0, 1, 1) in the recording codes (bk−i, . . . , and bk) setforth in Table 1 above.

In a state before the phase entrainment is completed, the differentialmetric detection portion 114 detects a phase error using the 36 statetransition patterns set forth in Table 1 above. This is because there isa need to increase a gain of the PLL loop as high as possible in orderto complete the phase entrainment sooner. Meanwhile, in a state afterthe phase entrainment is completed, the differential metric detectionportion 114 stops detecting a phase error using the state transitionpatterns relating to the 2 T pattern. The detection is stopped becauseextraction of the phase information from the 2 T mark is quite difficultat a recording density at which the frequency of the 2 T mark becomeshigher than the OTF cutoff frequency, the detection possibly becomes adisturbance on the contrary when the quality is poor. The phaseinformation is absent at each of the leading edge or the trailing edgeof the 2 T mark. However, according to the phase error extraction methodof this embodiment, it is possible to extract the phase information fromthe waveform in the amplitude direction obtained from intersymbolinterference with space lengths before and behind, and the detection maybe useful during the phase entrainment in some cases.

It should be appreciated, however, that this embodiment is not limitedto the switching control as described above. In a case where there is amargin of phase entrainment time, the phase error may be extractedalways without using state transition patters relating to the 2 Tpattern. Alternatively, in a case where it is known beforehand that therecording quality is satisfactory, the phase information may beextracted always from the state transition patterns relating to the 2 Tpattern. In addition, the example above described the control as to theON/OFF switching of a part of the state transition patterns set forth inTable 1 above. However, combinations of other state transition patternsset forth in Table 1 above may be used as well.

In addition, the above described an example of the frequency controlusing the synchronization pattern contained in the data. However, thefrequency control may use a track wobble frequency of a BD-RE/R.

An example of the phase error calculation and a manner to maintain thePLL loop gain constant of this embodiment will now be described.

For the 36 state transition patterns set forth in Table 1 above,concrete waveforms are shown in FIG. 3 and described in the above.

The phase error can be calculated, for example, in accordance withEquation (2):

$\begin{matrix}{{{PHASE}\mspace{14mu} {ERROR}} = \frac{{{{\sum\limits_{i = 0}^{4}( {{{Path}\; A_{i}} - S_{i}} )^{2}} - {\sum\limits_{i = 0}^{4}( {{{Path}\; B_{i}} - S_{i}} )^{2}}}} - 14}{14 \times 2}} & (2)\end{matrix}$

In Equation (2) above, the path A and the path B are PR equalizationideal values of five samples of the state transition patterns set forthin Table 1 above, and S is a reproduced waveform of five samplescorresponding to the predetermined state transition patterns set forthin Table 3 above. In the case of Table 1 above, because a squaredistance between ideal two state transition patterns is 14, thedetection window is 14×2. A code is appended to each state transitionpattern set forth in Table 1 above, that is, to each leading end andeach trailing end, so that it is detected as a phase error.

Regarding an amount of the phase error expressed by Equation (2) above,a value corresponding to the quality of a reproduced signal can becalculated by finding scattering (a variance) of the numerator andnormalizing the scattering with the detection window. In a case wherethe reproduction quality is poor, a variance in amount of the phaseerror detected in accordance with Equation (2) above becomes larger andin a case where the reproduction quality is satisfactory, a variance inamount of the phase error detected in accordance with Equation (2) abovebecomes smaller. In order to make the PLL loop gain independently of thereproduction quality to the extent possible, by lowering the gain of theLPF (not shown) included in the synchronization portion 115 and passingthe phase error in a case where the reproduction quality is poor and byincreasing the gain of the LPF (not shown) included in thesynchronization portion 115 and passing the phase error in a case wherethe reproduction quality is satisfactory, the same PLL loopcharacteristic can be achieved constantly.

In a case where an amount of the phase error is equal to or greater thanthe predetermined amount expressed by Equation (2) above, the phasedetection portion 112 may opt not to output the phase error to thesynchronization detection portion 115. More specifically, when thedetected phase error is greater than a predetermined threshold value,the phase detection portion 112 does not output the phase error to thesynchronization detection portion 115. Accordingly, it is understoodthat it is highly likely that a phase error equal to or greater than thepredetermined error is outputted erroneously. Hence, the PLL can bestabilized by deleting this phase error.

FIG. 8 is a view showing an example of a variance (frequencydistribution) of the phase error found in accordance with Equation (2)above. In the case of the phase detection using the state transitionpatterns with which the square of a Euclidean distance in Equation (2)above is 14, the detection width is ±14 with 0 at the center. The loopgain of the PLL is changed adequately according to the magnitude of avariance a of this distribution. As the changing method, for example,the loop gain may be changed linearly or in the shape of a quadraticcurve according to a change of a variance a.

A large phase error detected after the PLL lock state can be adisturbance to the PLL loop. It is therefore preferable to set apredetermined threshold value a for the phase error, so that the phaseerror is not outputted when the phase error is greater than thethreshold value ±α. An operation not to output the phase error means,for example, an operation to output 0. Because the abscissa of FIG. 8 isused for an amount of the phase error, in a case where the absolutevalue is greater than the predetermined threshold value ±α, it ispreferable not to output the phase error. For example, the predeterminedthreshold ±α may be ±7, which is half the detection width, ±14.

Alternatively, the loop gain may be changed adaptively so that a part ofthe detection patterns set forth in Table 1 above are selected accordingto a variance in amount of the phase error expressed by Equation (2)above so that the selected detection patterns are outputted. Forexample, in a case where the reproduction quality is satisfactory, thephase detection portion 112 may perform the phase detection using allthe statues transition patterns set forth in Table 1 above and in a casewhere the reproduction quality is poor, it may detect the phase bydeleting the state transition patterns relating to the 2 T pattern alonein Table 1 set forth above. Consequently, the PLL can be stabilized.Further, the phase detection portion 112 may be configured not tooutput, as the phase error, state transition patterns having aconsiderable variance among the state transition patterns set forth inTable 1 above.

The above described, as the method for detecting a phase error, a methodthat uses the differential metric information about the square of aEuclidean distance using the state transition patterns set forth inTable 1 above. Herein, a further devise to improve the accuracy in thephase detection for synchronization using Equation (2) above will bedescribed.

In order to overcome intersymbol interference and SNR deterioration inhigh density recording, Equation (2) above described a method ofdetecting the phase information from five points. It is true that thismethod is quite useful means. However, there is a case where the phaseinformation cannot be detected appropriately always, for example, whenthe input waveform is distorted significantly from the ideal waveform.As a case where the phase information cannot be detected appropriately,an asymmetric waveform of a recording mark and a space in which theinput waveform is distorted considerably is an outstanding example. Theasymmetric waveform is a waveform in which the maximum amplitude portionof a recording mark or the maximum amplitude portion of a spacedisplaces considerably from the ideal waveform. Another example is acase where the phase information cannot be detected appropriatelybecause information about a waveform displaced significantly from theideal waveform is detected as the phase information.

FIG. 9 is a view showing examples of an input waveform S in a case wherethe phase displaces significantly, an ideal waveform PA of the path A,and an ideal waveform PB of the path B. FIG. 9 shows an example in acase where the path A is detected as a correct path and the phase isdetected with a delay of about 90 degrees from the ideal point of thepath A. Also, examples of concrete numerical values of the inputwaveform S, the ideal waveform PA of the path A, and the ideal waveformPB of the path B are set forth in Table 5 below.

TABLE 5 SAMPLE [T] 0 1 2 3 4 INPUT WAVEFORM S 1.5 4 5.5 5.5 4.2 IDEALWAVEFORM PA OF PATH A 1 3 5 6 5 IDEAL WAVEFORM PB OF PATH B 0 1 3 4 4(PathA − S)² 0.25 1 0.25 0.25 0.64 (PathB − S)² 2.25 9 6.25 2.25 0.04

The phase error calculated in according with Equation (2) above usingthe numerical values set forth in Table 5 above is found to be +3.4.Because the phase error is calculated using sample values at five pointsin accordance with Equation (2) above, there is a case where distortionof the waveform at the sample time 0 and the sample time 4 hassignificant influences on the phase information. To avoid such aninconvenience, this embodiment proposes a method by which the sampletime 0 and the sample point 4 susceptible to distortion of the waveformamong the five points are not used for the phase error detection.

More specifically, the phase detection portion 112 extracts, during themaximum likelihood decoding, a phase error using partial responseequalization ideal values at sample times other than the top sample timeand the last sample time from the state transition patterns having onlya single zero cross point among the differential metrics at a pluralityof merging points at which a set of paths branched from a given statewill merge.

In this case, in Equation (2) above, only the sample times 1, 2, and 3are used. The phase error detection is thus expressed as Equation (3):

$\begin{matrix}{{{PHASE}\mspace{14mu} {ERROR}} = {{{{\sum\limits_{i = 1}^{3}( {{{Path}\; A_{i}} - S_{i}} )^{2}} - {\sum\limits_{i = 1}^{3}( {{{Path}\; B_{i}} - S_{i}} )^{2}}}} - 12}} & (3)\end{matrix}$

The phase error calculated using Equation (3) above is found to be +4.0.Hence, the phase displacement can be detected at a higher degree ofaccuracy than in accordance with Equation (2) above. Hence, even in acase where the phase is distorted significantly it is still possible todetect the phase error at a high degree of accuracy by detecting thephase error in accordance with Equation (3) above. It is thereforepossible to increase the gain of the phase loop appropriately.Consequently, it is expected that a rate and stability of the phaseentrainment of the PLL can be enhanced.

Herein, attention has been paid to the state transition patterns setforth in Table 1 above and in which the square of a Euclidean distanceis 14 as the patterns in which the phase error by PR 12221 ML can bedetected. Further, by also taking distortion of the waveform intoaccount, it has been proposed not to detect the phase information (errorinformation) from all the ideal waveforms of five samples, but to detectthe phase error from the sample values less susceptible to distortion ofthe waveform. Owing to the proposals of this embodiment, it becomespossible to provide the most suitable phase detection portion for thewaveforms having undergone the influences of intersymbol interference,SNR deterioration, and the waveform distortion in the case of a highrecording density waveform.

With the phase detection portion of this embodiment, PR 12221 ML hasbeen described as a concrete example and a proposal has been made as tothe selection of sample values among the state transition patterns inwhich the square of a Euclidean distance is 14. However, when anothertype of PRML is used, attention should be paid to the squares of otherEuclidean distances, so that the phase detection is performed on thebasis of the sample values less susceptible to various stresses in thestate transition patterns in which the square of a Euclidean distance isanother value. It should be appreciated that the sample values used whendetecting a phase error is not limited to the three sample valuesspecified above in this embodiment. The phase error only has to bedetected from appropriate sample points in PRML adopted, and the phaseerror can be expressed by Equation (4):

$\begin{matrix}{{{PHASE}\mspace{14mu} {ERROR}} = {{{{\sum\limits_{i = n}^{m}( {{{Path}\; A_{i}} - S_{i}} )^{2}} - {\sum\limits_{i = n}^{m}( {{{Path}\; B_{i}} - S_{i}} )^{2}}}} - L}} & (4)\end{matrix}$

In Equation (4) above, n and m mean that the phase error can be detectedwith arbitrary samples among patterns with which the phase error can bedetected. Also, L is the square of a Euclidean distance between theideal waveform PA of the path A and the ideal waveform PB of the path Bfound from the samples determined with n and m.

More specifically, the phase detection portion 112 extracts, during themaximum likelihood decoding, a phase error using a PR (Partial Response)equalization ideal value at a second sample time less susceptible todistortion of the waveform of an input signal than at a first sampletime among the state transition patterns having a single zero crosspoint alone among the differential metrics at a plurality of mergingpoints at which a set of paths branched from a given state will merge.Herein, the first sample time includes the top sample time and the lastsample time, and the second sample time includes sample times other thanthe top sample time and the last sample time.

The configuration of a waveform shaping portion different from thewaveform shaping portion 108 shown in FIG. 7 will now be described. FIG.10 is a block diagram showing the configuration of the waveform shapingportion using a frequency sampling algorithm. FIG. 7 shows an example ofthe configuration in a case where the LMS algorithm is used. Withreference to FIG. 10, an example of the configuration to update thefilter coefficients adaptively using the frequency sampling algorithmwill be described.

Referring to FIG. 10, the waveform shaping portion 108 includes a PRequalization portion 109 and an adaptive coefficient update portion 110.The PR equalization portion 109 includes an FIR filter portion 201. TheFIR filter portion 201 in FIG. 10 is of the same configuration as theFIR filter portion 201 in FIG. 7. The adaptive coefficient updateportion 110 is formed of an amplitude detection portion 301, an LPF (LowPass Filter) 202, and a coefficient computation portion 303.

The amplitude detection portion 301 includes a binarization portion 220,a rising edge detection portion 221, a falling edge detection portion222, a first amplitude level extraction portion 223, a second amplitudelevel extraction portion 224, a third amplitude level extraction portion225, and a fourth amplitude level extraction portion 226. The LPF 302includes a first LPF 227, a second LPF 228, a third LPF 229, and afourth LPF 230. The coefficient computation portion 303 includes a firstamplitude level calculation portion 231, a second amplitude levelcalculation portion 232, a first amplitude error detection portion 233,a second amplitude error detection portion 234, an IDFT (InverseDiscrete Fourier Transform) portion 235, and a coefficient calculationportion 236.

The first amplitude level detection portion 211 includes the rising edgedetection portion 221, the falling edge detection portion 222, the firstamplitude level extraction portion 223, the second amplitude levelextraction portion 224, the first LPF 227, the second LPF 228, and thefirst amplitude level calculation portion 231. The second amplitudelevel detection portion 212 includes the rising edge detection portion221, the falling edge detection portion 222, the third amplitude levelextraction portion 225, the fourth amplitude level extraction portion226, the third LPF 229, the fourth LPF 230, and the second amplitudelevel calculation portion 232. The coefficient calculation portion 215includes the IDFT portion 235 and the coefficient calculation portion236.

The first amplitude level detection portion 211 detects an N′thamplitude level from the predetermined reference level in an output ofthe FIR filter portion 201. The second amplitude level detection portion212 detects an M′th (M>N) amplitude level from the predeterminedreference level in an output of the FIR filter portion 201.

The binarization portion 220 converts a digital filter output signaloutputted from the FIR filter portion 201 into a binary form. Thedigital filter output signal converted into a binary form is outputtedto the rising edge detection portion 221 and the falling edge detectionportion 222.

The rising edge detection portion 221 detects a rising edge (zero crosspoint) from the digital filter output signal outputted from the FIRfilter portion 201. The falling edge detection portion 222 detects afalling edge (zero cross point) from the digital filter output signaloutputted from the FIR filter portion 201.

The first amplitude level extraction portion 223 extracts an N′thamplitude level from the rising edge (zero cross point) detected by therising edge detection portion 221. It should be noted that, in thisembodiment, the first amplitude level extraction portion 223 extractsthe amplitude level following the rising edge detected by the risingedge detection portion 221.

The second amplitude level extraction portion 224 extracts an N′thamplitude level from the falling edge (zero cross point) detected by thefalling edge detection portion 222. It should be noted that, in thisembodiment, the second amplitude level extraction portion 224 extractsthe amplitude level following the falling edge detected by the fallingedge detection portion 222.

The third amplitude level extraction portion 225 extracts an M′th (M>N)amplitude level from the rising edge (zero cross point) detected by therising edge detection portion 221. It should be noted that, in thisembodiment, the third amplitude level extraction portion 225 extractsthe amplitude level, which is the amplitude level that follows theamplitude level following the rising edge (zero cross point) detected bythe rising edge detection portion 221 and monotonously increases fromthe amplitude level following the zero cross point.

The fourth amplitude level extraction portion 226 extracts an M′th (M>N)amplitude level from the falling edge (zero cross point) detected by therising edge detection portion 222. It should be noted that, in thisembodiment, the fourth amplitude level extraction portion 226 extractsthe amplitude level, which is the amplitude level that follows theamplitude level following the rising edge (zero cross point) detected bythe falling edge detection portion 222 and monotonously decreases fromthe amplitude level following the zero cross point.

The first LPF 227 passes the N′th amplitude level extracted by the firstamplitude level extraction portion 223. The second LPF 228 passes theN′th amplitude level extracted by the second amplitude level extractionportion 224.

The third LPF 229 passes the M′th amplitude level extracted by the thirdamplitude level extraction portion 225. The fourth LPF 230 passes theM′th amplitude level extracted by the fourth amplitude level extractionportion 226.

The first amplitude level calculation portion 231 calculates adifference (amplitude La) between the N′th amplitude level extracted bythe first amplitude level extraction portion 223 and the N′th amplitudelevel extracted by the second amplitude level extraction portion 224.

The second amplitude level calculation portion 232 calculates adifference (amplitude Lb) between the M′th amplitude level extracted bythe third amplitude level extraction portion 225 and the M′th amplitudelevel extracted by the fourth amplitude level extraction portion 226.

The first amplitude error detection portion 233 detects a differencebetween the amplitude level detected by the first amplitude leveldetection portion 211 and a first target amplitude. To be more concrete,the first amplitude error detection portion 233 calculates an amplitudeerror Aerr on the basis of the amplitude La calculated by the firstamplitude level calculation portion 231 and the first target amplitudeTa.

The second amplitude error detection portion 234 detects a differencebetween the amplitude level detected by the second amplitude leveldetection portion 212 and a second target amplitude that is differentfrom the first target amplitude. To be more concrete, the secondamplitude error detection portion 234 calculates an amplitude error Berron the basis of the amplitude Lb calculated by the second amplitudelevel calculation portion 232 and the second target amplitude Tb.

The coefficient calculation portion 215 calculates coefficients of theFIR filter portion 201 on the basis of the outputs of the firstamplitude error detection portion 233 and the second amplitude errordetection portion 234.

The IDFT portion 235 applies inverse discrete Fourier transform to theoutputs of the first amplitude error detection portion 233 and thesecond amplitude error detection portion 234. The coefficientcalculation portion 236 calculates the coefficients (tap coefficients C0through C4) of the FIR filter portion 201. A calculation method of thecoefficients will be described below.

Hereinafter, a concrete operation of the waveform shaping portion 108shown in FIG. 10 will be described.

The frequency sampling algorithm is a method of designing thecoefficients of an FIR filter by which an impulse response h[n] (filtercoefficient) is found by applying inverse discrete Fourier transform(IDFT) to sample values found by sampling the frequency response between−ωs/2 to ωs/2 at N points (the frequency width is ωs/N). It ischaracterized in that, in this embodiment, in order to find the filtercoefficients adaptively, the respective band gains are fed back so thatthe amplitude of the band becomes the target amplitude, and the FIRfilter coefficients are controlled using the frequency samplingalgorithm on the basis of the respective band gains that have been fedback. In general, the impulse response h[n] is expressed by Equation (5)and Equation (6):

$\begin{matrix}{{h\lbrack n\rbrack} = {\frac{1}{N}{\sum\limits_{k = {{{- N}/2} + 1}}^{N/2}{{H(k)}W^{- {kn}}}}}} & (5) \\{W = ^{{- j}\frac{2}{N}}} & (6)\end{matrix}$

An example of the method of finding the coefficients of a 9-tap FIRfilter (digital equalizer) will now be described using FIG. 11, FIG. 12,and FIG. 13. FIG. 11 is a view showing the gain characteristic targetvalues at the respective frequencies of the 9-tap FIR filter (digitalequalizer). FIG. 12 is a view showing the tap coefficients (C0 throughC4) of the digital equalizer found in accordance with Equation (5) aboveusing the frequency sampling algorithm on the basis of the gaincharacteristic target values. FIG. 13 is a view showing the frequencycharacteristic of the digital equalizer calculated on the basis of thetap coefficients.

The amplitude detection portion 301 detects the amplitude of thewaveform that has been passed through the digital equalizer having thecharacteristics shown in FIG. 11, FIG. 12, and FIG. 13 and feeds backthe detection result to the coefficients of the nine taps so that thedetection result becomes the target amplitudes at the respectivefrequency bands. A description will be given in an example case where aBD is reproduced at 1× speed as follows. That is, the frequency samplinginterval to control the gain is found to be: 66 MHz (9 taps+1)=6.6 MHz.Herein, as is shown in FIG. 11, by setting the gain characteristictarget values at an interval of about 6.6 MHz, points A0 through A5 canbe defined. The point A0 is a gain of DC (0 MHz), the point A1 is a gainof 5 T (6.6 MHz), the point A2 is a gain of 2.5 T (13.2 MHz), the pointA3 is a gain of 1.67 T (19.8 MHz), the point A4 is a gain of 1.25 T(26.4 MHz), and the point A5 is a gain of 1 T (33.0 MHz).

The tap coefficients C0 through C4 of the digital equalizer calculatedin accordance with Equation (5) above are found, respectively, byEquations (7) through (11):

$\begin{matrix}{C_{0} = {\frac{A_{0}}{10} + \frac{A_{1} + A_{2} + A_{3} + A_{4}}{5}}} & (7) \\{C_{1} = {\frac{A_{0}}{10} + {0.162A_{1}} + {0.062A_{2}} - {0.062A_{3}} - {0.162A_{4}}}} & (8) \\{C_{2} = {\frac{A_{0}}{10} + {0.062A_{1}} - {0.162A_{2}} - {0.162A_{3}} + {0.062A_{4}}}} & (9) \\{C_{3} = {\frac{A_{0}}{10} - {0.062A_{1}} - {0.162A_{2}} + {0.162A_{3}} + {0.062A_{4}}}} & (10) \\{C_{4} = {\frac{A_{0}}{10} - {0.162A_{1}} + {0.062A_{2}} + {0.062A_{3}} - {0.162A_{4}}}} & (11)\end{matrix}$

The amplitude detection method and the feedback control on the band gainwill now be described. FIG. 14A is a view used to describe the amplitudedetection in a coefficient calculation using the frequency samplingalgorithm. FIG. 14B is a view showing the processing result by the LPFat the respective amplitude levels A through D shown in FIG. 14A.

The amplitude detection portion 301 detects, in reference to the zerocross point of an output of the digital equalizer, a sample pointfollowing the zero cross point and a sample point that follows thesample point following the zero cross point and monotonously increasesor monotonously decreases from the sample point following the zero crosspoint.

In the case of the amplitude level A or the amplitude level C shown inFIG. 14A, the sample point following the zero cross point is detected asan average amplitude of 2 T and 3 T. In the case of the amplitude levelB or the amplitude level D shown in FIG. 14A, the sample point thatfollows the sample point following the zero cross point and monotonouslyincreases or monotonously decreases from the sample point following thezero cross point is detected as an average amplitude of 3 T and 4 T.Each sample point is detected in the amplitude detection portion 301shown in FIG. 10.

FIG. 14B shows a manner in which the amplitudes detected at therespective amplitude levels A through D of FIG. 14A are passed throughthe LPF having the predetermined characteristic. An amplitude error whenthe amplitude level A and the amplitude level C are passed through theLPF can be detected as the amplitude La in about the 2.5 T band.Likewise, an amplitude error when the amplitude level B and theamplitude level D are passed through the LPF can be detected as theamplitude Lb in about the 3.5 T band.

The coefficient computation portion 303 calculates an amplitude error onthe basis of the amplitude La and the amplitude Lb in the 2.5 T band andthe 3.5 T band, respectively, and the target amplitude Ta and the targetamplitude Tb. In other words, the coefficient computation portion 303calculates an amplitude error, Aerr=Ta−La, corresponding to the 2.5 Tband, and an amplitude error, Berr=Tb−Lb, corresponding to the 3.5 Tband.

Subsequently, the coefficient computation portion 303 finds A0 throughA4 by adding up the amplitude error Aerr and the amplitude Berr and addsthe result to the respective band gains. For example, the coefficientcomputation portion 303 is able to find A0 through A4 as follows: A0=afixed gain of one; A1=a gain of one+Berr; A2=a gain of one+Aerr; A3=again of one+Aerr; and A4=a fixed gain of one or less. Once A0 through A4are found, the coefficient computation portion 303 calculates the tapcoefficients C0 through C4 using Equations (7) through Equations (11)above, respectively. The coefficient computation portion 303 thenupdates the current tap coefficients in the FIR filter portions 201 tothe tap coefficients C0 through C4 calculated as above.

In this manner, it is possible to update the tap coefficients of thedigital equalizer adaptively in response to the amplitude of an inputwaveform. The amplitude error and the tap coefficients described aboveare calculated in the LPF 302 and the coefficient computation portion303 in FIG. 10.

The wave shaping portion 108 inserted in the loop of the PLL isnecessarily formed of an adaptive coefficient update filter.Accordingly, the coefficients calculated using the frequency samplingalgorithm are controlled to be right-left symmetrical. The coefficientcontrol under the restriction of right-left symmetry is the control tochange a gain at a predetermined frequency alone without having tochange the phase relation between the digital reproduced signal inputtedinto the FIR filter portion 201 and the digital filter output signaloutputted therefrom.

In FIG. 14, the amplitude level A and the amplitude level B upper thanthe reference and the amplitude level C and the amplitude level D lowerthan the reference are detected, and a difference between the amplitudelevel A and the amplitude level C is defined as the 2.5 T band amplitudeand a difference between the amplitude level B and the amplitude level Dis defined as the 3.5 T band amplitude. The invention, however, is notlimited to this configuration. Given that the amplitude level A upperthan the reference and the amplitude level C lower than the referenceare almost the same and the amplitude level B upper than the referenceand the amplitude level D lower than the reference are almost the same,it is sufficient to detect two amplitudes.

The embodiment using FIG. 11 is of the configuration to detect aplurality of amplitudes. The invention, however, is not limited to thisconfiguration. It may be configured in such a manner so as to detectamplitudes of particular frequencies alone. Even when only one amplitudeis to be detected, adequate coefficients can be calculated given thatthe amplitudes of a mark and a space are the same or on the preconditionthat the amplitudes of a mark and a space are the same. For example, ofA0 through A5 described above, the amplitude of A3 alone may be detectedand the amplitudes of the rest of A0, A1, A2, A4, and A5 may beestimated from the amplitude of A3. The result may possibly vary from adesired filter characteristic in comparison with a case where theamplitude values of all A0 through A5 are detected. However, adetermination can be made from the selection of the circuit size and theperformance. This alternative method is more useful in a case where afilter having fewer taps is used.

The adaptive coefficient update filter is inserted in the loop of thePLL of this embodiment. The purpose of this configuration is to meet theneed for a waveform that will be inputted into the maximum likelihooddecoding portion 111 to be shaped into the waveform expected in Viterbidecoding before it is inputted because the maximum likelihood decodingportion 111 extracts a phase error during the Viterbi decoding process.By inputting a waveform after it is shaped into the waveform expected inViterbi decoding, phase error detection accuracy can be enhanced, whichmakes the PLL control more stable.

The adaptive coefficient update filter using the frequency samplingalgorithm does not require binarization accuracy that is necessary togenerate a teacher signal (expected value level) in the LMS algorithm.The adaptive coefficient update filter using the frequency samplingalgorithm is therefore able to calculate adequate filter coefficients ina stable manner for a poor quality waveform.

In addition, it is possible to achieve stable adaptive control evenduring the PLL entrainment (non-synchronous state).

The adaptive coefficient update filter using the LMS algorithm finds therespective tap coefficients from the correlation with a waveforminputted therein. The adaptive coefficient update filter using the LMSalgorithm therefore has a high degree of freedom in controlling therespective tap coefficients in comparison with the frequency samplingalgorithm and is thus able to filter the waveform to have thecharacteristic closer to the expected PR characteristic.

Although it is not shown in FIG. 6, a block that appropriately controlsthe DC level of a waveform may be inserted between the A/D conversionportion 106 and the waveform shaping portion 108. A waveform to beinputted into the A/D conversion portion 106 is controlled to beamplitude-centered by the AGC/OFFSET control portion 107 so that it canfully use the D range. The block that appropriately controls the DClevel controls the DC level to be energy-centered and inputs the resultinto the waveform shaping portion 108. The control on the DC level isperformed particularly so as to apply filtering to an asymmetricwaveform in a more appropriate manner. It should be noted, however, thateven when the block that appropriately controls the DC level isinserted, it is still necessary to input the waveform into theAGC/OFFSET control portion 107 from the A/D conversion portion 106.

It should be appreciated that the adaptive coefficient update waveformshaping method using the frequency sampling algorithm of this embodimentis not a method corresponding only to a reproduced waveform in a casewhere an optical disc having a higher recording line density than aconventional optical disc is to be reproduced. It is also applicable toa reproduced waveform of a conventional BD having a recording capacityof 25 GB as well as to a reproduced waveform of a DVD or a CD. Theequalizer gains only have to be set to achieve the desiredcharacteristic.

The embodiment above described the adaptive coefficient update methodusing the frequency sampling algorithm in a case where there are ninetaps. It should be appreciated, however, that the tap coefficients canbe updated adaptively in the same manner as above even there are adifferent number of taps. For example, the FIR filter portion 201 mayhave seven taps or eleven taps.

Second Embodiment

An optical disc device according to a second embodiment of the inventionwill be described first. FIG. 15 is a block diagram showing theconfiguration of the optical disc device according to the secondembodiment of the invention.

The optical disc device of FIG. 15 is almost of the same configurationand operates almost in the same manner as the optical disc device ofFIG. 6. Accordingly, only differences in configuration and operationfrom the optical disc device of FIG. 6 will be described herein.

In an optical disk device 200 shown in FIG. 15, the waveform shapingportion 108 includes a first adaptive coefficient update portion 110Aand a second adaptive coefficient update portion 110B. The firstadaptive coefficient update portion 110A updates the tap coefficients ofthe FIR filter in the PR equalization portion 109 using the LMSalgorithm. The second adaptive coefficient update portion 110B updatesthe tap coefficients of the FIR filter in the PR equalization portion109 using the frequency sampling algorithm.

The adaptive coefficient update filter using the frequency samplingalgorithm does not require binarization accuracy that is necessary togenerate a teacher signal (expected value level) in the LMS algorithm.Hence, the adaptive coefficient update filter using the frequencysampling algorithm is able to calculate adequate filter coefficients fora poor quality waveform in a stable manner. It is also possible toachieve stable adaptive control during the PLL entrainment (asynchronousstate).

The adaptive coefficient update filter using the LMS algorithm findsrespective tap coefficients from the correlation with a waveforminputted therein. The adaptive coefficient update filter using the LMSalgorithm therefore has a higher degree of freedom in controlling therespective tap coefficients in comparison with the frequency samplingalgorithm and is thus able to filter the waveform to have acharacteristic closer to the expected PR characteristic.

In this embodiment, two adaptive coefficient update portions areswitched in response to a state of the optical disc device by utilizingthe characteristics of these two adaptive equalization methods, that is,the frequency sampling algorithm and the LMS algorithm. By appropriatelyswitching the two adaptive coefficient update portions, the optical discdevice is able to perform an optimal gain correction that best suits aninput waveform through selection of the adaptive method that best suitsthe situation. Consequently, not only is it possible to stabilize thePLL control, but it is also possible to enhance the convergence of thePLL control (shortening of the entrainment time). Because theconfigurations of the respective adaptive coefficient update portionshave been described in the first embodiment above, the description isomitted herein.

An operation of the optical disc device configured as above will now bedescribed.

The synchronization entrainment control portion 117 in thesynchronization detection portion 115 controls, as is set forth in Table6 below, the AGC/OFFSET control portion 107, the first adaptivecoefficient update portion 110A in the waveform shaping portion 108, thesecond adaptive coefficient update portion 110B in the waveform shapingportion 108, and the particular pattern detection portion 113 in thephase detection portion 112.

TABLE 6 FIRST ADAPTIVE SECOND ADAPTIVE AGC/OFFSET COEFFICIENTCOEFFICIENT CONTROL UPDATE UPDATE PARTICULAR PATTERN CONTROL STATEPORTION 107 PORTION 110A PORTION 110B DETECTION PORTION 113 BEFORETRACKING OFF OFF OFF OFF CONTROL IS ENABLED BEFORE AGC/OFFSET ON OFF ONOFF CONVERGENCE IS ENABLED BEFORE FREQUENCY ON OFF ON OFF (DETECTENTRAINMENT IS ENABLED SYNCHRONIZATION MARK) BEFORE PHASE ON OFF ONTABLE 1 ENTRAINMENT IS ENABLED (ALL PATTERNS) AFTER PHASE ON ON OFFTABLE 1 ENTRAINMENT IS ENABLED (DELETE 2T PATTERNS)

As examples of the control state, states are classified into statesbefore the tracking control is enabled, before the AGC/OFFSETconvergence is enabled, before the frequency entrainment is enabled,before the phase entrainment is enabled, and after the phase entrainmentis enabled, each of which will be described below.

The content of Table 6 above is almost the same as the operation of theoptical disc device described using Table 4 above in the firstembodiment above. Herein, a description will be given only to a portioninvolved with the switching control operation on the adaptivecoefficient update portions in the waveform shaping portion 108, whichis different from the content of Table 4 above.

In a state before the tracking control is enabled, the synchronizationentrainment control portion 117 in the synchronization detection portion115 maintains the control on the first adaptive coefficient updateportion 110A and the second adaptive coefficient update portion 110B inthe waveform shaping portion 108 in an OFF state.

In a state after the tracking control is enabled, the synchronizationentrainment control portion 117 in the synchronization detection portion115 activates the second adaptive coefficient update portion 110B in thewaveform shaping portion 108 and keeps it operating until the phaseentrainment is completed. Meanwhile, the PR equalization portion 109receives tap coefficients that will update the current ones constantlyfrom the second adaptive coefficient update portion 110B.

In a state after the phase entrainment is enabled, the synchronizationentrainment control portion 117 in the synchronization detection portion115 switches the operation control from the second adaptive coefficientupdate portion 110B to the first adaptive coefficient update portion110A in the waveform shaping portion 108. The PR equalization portion109 thus receives the tap coefficients that will update the current onesconstantly from the first adaptive coefficient update portion 110A. Inthis instance, the first adaptive coefficient update portion 110A usesthe coefficients calculated in the second adaptive coefficient updateportion 110B as the initial values of the tap coefficients when thecontrol is switched to the first adaptive coefficient update portion110A.

Table 6 above shows a case where the AGC/OFFSET control portion 107 andthe second adaptive coefficient update portion 110B in the waveformshaping portion 108 are operated simultaneously after the trackingcontrol is enabled. The invention, however, is not particularly limitedto this configuration. It may be configured in such a manner that theAGC/OFFSET control portion 107 alone is activated first after thetracking control is enabled and then the second adaptive coefficientupdate portion 110B in the waveform shaping portion 108 is activatedwhen an operation of the AGC/OFFSET portion 104 becomes closer to thepredetermined target value.

This is because both the first adaptive coefficient update portion 110Aand the second adaptive coefficient update portion 110B in the waveformshaping portion 108 are designed to operate in a stable manner on theprecondition that the offset and the amplitude of the waveform fallwithin a predetermined value. In addition, because the adaptivecoefficient update control also has a capability of correcting theamplitude of the waveform, it possibly becomes unstable in associationwith the AGC that corrects the amplitude of the waveform.

It is therefore preferable to perform control by clearly defining thecapabilities and the bands. Accordingly, it is more preferable that thesynchronization entrainment control portion 117 does not activate thesecond adaptive coefficient update portion 110B or the phase detectionportion 112 until the operation of the AGC/OFFSET portion 104 convergeswithin the predetermined value. The phrase, “to fall within thepredetermined value”, referred to herein means a case where the AGCcontrol or the OFFSET control has converged to a range within about 90%of the target value. For example, given 800 mV as the target ACamplitude, then it is sufficient for the AGC control to reach 720 mV orgreater. The same can be said about the OFFSET control. Thepredetermined value can be set to a value at which the system isstabilized.

It should be appreciated that this embodiment is not limited to theexamples of the control states set forth in Table 6 above. In othercontrol states and in the control states set forth in Table 6 above, thecontrol only has to be optimized so that the system is stabilized.

The embodiment above described the phase detecting device, the opticaldisc device, and the phase detection method for the synchronizationmethod of a reproduced signal in a case where the recording line densityof an optical disc is enhanced. Herein, influences of the enhancedrecording line density on the waveform will be described using FIG. 16and FIG. 17 in the case of a BD by way of example.

FIG. 16 is a view showing the relative relation between a mark sequencerecorded on a track of an optical disc and a light beam diameter. As ina DVD, recording data is recorded in a BD in the form of mark sequencesas a physical change on an optical disc. A mark sequence having theshortest length among the mark sequences is referred to the shortestmark. In the case of a BD having a recording capacity of 25 GB, thephysical length of the shortest mark 402 is 0.149 μm. The shortest marklength of a BD corresponds about 1/2.7 of the shortest mark length of aDVD. It is therefore nearing the limit of the optical resolution atwhich a recording mark can be identified by a light beam even when theresolution of the laser is increased by changing the wavelengthparameter (405 nm) and the NA parameter (0.85) in the optical system.

Referring to FIG. 16, a light beam 403 is irradiated to a mark sequencerecorded on a track 401. In a BD, the diameter of a light spot is about0.39 μm because of the optical system parameters specified above. In acase where the recording line density is enhanced without changing thestructure of the optical system, the recording mark becomes smallerrelatively with respect to the light spot diameter, which deterioratesthe reproduction resolution.

In a case where a recording mark is reproduced with a light beam, theamplitude of a reproduced signal decreases as the recording mark becomesshorter and reaches 0 at the limit of the optical resolution. Theinverse number of the cycle of the recording mark is referred to as aspatial frequency, and the relation between the spatial frequency andthe signal amplitude is referred to as an OTF (Optical TransferFunction). The signal amplitude decreases almost linearly as the spatialfrequency becomes higher and the limit of reproduction at which thesignal amplitude decreases to 0 is referred to as the OTF cutoff.

FIG. 17 is a view showing the OTF of a BD having a recording capacity of25 GB. The spatial frequency of the shortest mark of a BD is about 80%of the OTF cutoff and nears the OTF cutoff. Also, it is understood thatthe reproduction amplitude of the shortest mark becomes as small asabout 10%. In the case of a BD, when a recording capacity is as large asabout 31 GB, the frequency of the shortest mark is the OTF cutoff andthe reproduction amplitude is hardly detected. When the frequency of theshortest mark is in the vicinity of the OTF cutoff frequency or exceedsthe OTF cutoff frequency, the reproduction amplitude of a reproducedsignal becomes small because the resolution of the laser has reached orexceeded its limit. The SNR therefore deteriorates abruptly.

As has been described, the relation between the spatial frequency andthe signal amplitude is defined by the OTF. The signal amplitudedecreases almost linearly as the spatial frequency becomes higher andthe limit of reproduction at which the signal amplitude decreases to 0is defined as the cutoff frequency of the OTF. The shortest markfrequency of a reproduced signal recorded in an information recordingmedium is in the vicinity of the cutoff frequency of the OTF.

In this embodiment, the phrase, “in the vicinity of the OTF cutofffrequency”, specifies a range from the 2 T frequency to the OTF cutofffrequency within which a recording capacity of a BD per layer is 30 GB.In this case, the physical length of 2 T is about 0.124 μm. In short,the frequency of the shortest mark is in the vicinity of the OTF cutofffrequency. Also, the phrase, “a frequency exceeding the OTF cutofffrequency”, means a density at which recoding is performed with a lengthnot longer than about 0.124 μm, which is the physical length of 2 T.

This embodiment is to propose a synchronous detection method in a casewhere a region in which information is recorded at a recording linedensity at which the shortest mark frequency of a reproduced signalrecorded in an optical disc is in the vicinity of the OTF cutofffrequency or exceeds the OTF cutoff frequency.

To be more specific, in order to perform the binarization process andthe signal evaluation process, for example, a waveform shaping portionand a maximum likelihood decoding portion for the PRML process may beprovided in the latter stage of the waveform shaping portion 108 in FIG.6 and FIG. 15 apart from those of the PRML process for synchronizationdetection of this embodiment.

The components of the optical disc device of this embodiment can beachieved in the form of an LSI (Large Scale Integration) that is anintegrated circuit. The components provided to the optical disc devicemay be formed in such a manner that each component is formed in one chipor the components are integrated into one chip either partially orentirely.

The integrated circuit is referred to as an LSI herein. It should benoted, however, that it may be referred to also as an IC (IntegratedCircuit), an LSI, a super LSI, or an ultra-LSI because of a differencein integration degree.

Also, it should be appreciated that the integrated circuit of thisembodiment is not limited to an LSI and it may be achieved by anexclusive-use circuit or a general-purpose processor. Also, it is alsopossible to use an FPGA (Field Programmable Gate Array) that can beprogrammed after the LSI is fabricated or a reconfigurable processor inwhich the connections and settings of circuit cells inside the LSI arereconfigurable.

Further, when a circuit integration technique that replaces an LSIbecomes available owing the advancement in the semiconductor technologyor other derivative techniques, it goes without saying that thefunctional blocks can be integrated using this technique. Thebiotechnology has a potential of such applications.

The specific embodiments described above chiefly contain inventionshaving the following configurations.

A phase error detecting device according to an aspect of the inventionincludes: a waveform shaping portion that receives a digital reproducedsignal generated from an analog reproduced signal reproduced from aninformation recording medium and shapes a waveform of the digitalreproduced signal; a maximum likelihood decoding portion that appliesmaximum likelihood decoding to the digital reproduced signal in thewaveform shaped by the waveform shaping portion and generates abinarized signal indicating a result of the maximum likelihood decoding;a phase detection portion that detects a phase error on the basis of thedigital reproduced signal in the waveform shaped by the waveform shapingportion and the binarized signal generated by the maximum likelihooddecoding portion; and a synchronization detection portion that generatesa reproduction clock signal using the phase error detected by the phasedetection portion and brings the digital reproduced signal insynchronization with the reproduction clock signal that has beengenerated. The phase detection portion extracts, during the maximumlikelihood decoding, the phase error using state transition patternshaving only a single zero cross point among differential metrics at aplurality of merging points at which a set of paths branched from agiven state merges.

According to this configuration, a digital reproduced signal generatedfrom an analog reproduced signal reproduced from an informationrecording medium is received and the waveform of the digital reproducedsignal is shaped by the waveform shaping portion. The digital reproducedsignal in the shaped waveform is then subjected to maximum likelihooddecoding by the maximum likelihood decoding portion. A binarized signalindicating the result of the maximum likelihood decoding is thusgenerated by the maximum likelihood decoding portion. Thereafter, aphase error is detected by the phase detection portion on the basis ofthe digital reproduced signal in the shaped waveform and the binarizedsignal that has been generated. In this instance, the phase error isextracted during the maximum likelihood decoding using the statetransition patterns having only a single zero cross point among thedifferential metrics at a plurality of merging points at which a set ofpaths branched from a given state merges. A reproduction clock signal isthen generated using the phase error that has been detected and thedigital reproduced signal is brought into synchronization with thereproduction clock signal that has been generated by the synchronizationdetection portion.

Accordingly, the phase error is extracted during the maximum likelihooddecoding using the state transition patterns having only a single zerocross point among the differential metrics at a plurality of mergingpoints at which a set of paths branched from a given state merges.Accuracy in detecting the phase error can be therefore enhanced, whichmakes it possible to generate a reproduction clock signal in a stablemanner.

Also, it is preferable for the phase error detecting device describedabove that a relation between a spatial frequency and a signal amplitudeis defined by an optical transfer function, the signal amplitudedecreases almost linearly as the spatial frequency becomes higher, and alimit of reproduction at which the signal amplitude decreases to 0 isdefined as a cutoff frequency of the optical transfer function, then ashortest mark frequency of a reproduced signal recorded in theinformation recording medium is in a vicinity of the cutoff frequency ofthe optical transfer function.

According to this configuration, the relation between the spatialfrequency and the signal amplitude is defined by the optical transferfunction. In the optical transfer function, the signal amplitudedecreases almost linearly as the spatial frequency becomes higher. Thelimit of reproduction at which the signal amplitude decreases to 0 isdefined as the cutoff frequency of the optical transfer function. Then,the shortest mark frequency of a reproduced signal recorded in theinformation recording medium is in the vicinity of the cutoff frequencyof the optical transfer function.

It is therefore possible to reproduce an optical disc in whichinformation is recorded at a recording line density at which theshortest mark frequency is in the vicinity of the cutoff frequency ofthe optical transfer function.

Also, it is preferable for the phase error detecting device describedabove that the phase detection portion extracts, during the maximumlikelihood decoding, the phase error using a partial responseequalization ideal value at a second sample time at which influences ofdistortion of a waveform of an input signal are less than at a firstsample time from the state transition patterns having only a single zerocross point among the differential metrics at a plurality of mergingpoints at which a set of paths branched from a given state merges.

According to this configuration, the phase error is extracted during themaximum likelihood decoding using a partial response equalization idealvalue at the second sample time at which influences of distortion of thewaveform of an input signal are less than at the first sample time fromthe state transition patterns having only a single zero cross pointamong the differential metrics at a plurality of merging points at whicha set of paths branched from a given state merges.

Accordingly, by extracting the phase error using the partial responseequalization ideal value at the second sample time at which theinfluences of distortion of the waveform of an input signal are lessthan at the first sample time, it becomes possible to detect the phaseerror at satisfactory accuracy even when the phase displacessignificantly.

Also, it is preferable for the phase error detecting device describedabove that: the first sample time includes a top sample time and a lastsample time; the second sample time includes a sample time other thanthe top sample time and the last sample time; and the phase detectionportion extracts, during the maximum likelihood decoding, the phaseerror using the partial response equalization ideal value at a sampletime other than the top sample time and the last sample time from thestate transition patterns having only a single zero cross point amongthe differential metrics at a plurality of merging points at which a setof paths branched from a given state merges.

According to this configuration, the phase error is extracted during themaximum likelihood decoding using a partial response equalization idealvalue at a sample time other than the top sample time and the lastsample time from the state transition patterns having only a single zerocross point among the differential metrics at a plurality of mergingpoints at which a set of paths branched from a given state merges.

The partial response equalization ideal values at the top sample timeand the last sample time are susceptible to distortion of the waveformof an input signal. Hence, by detecting the phase error using thepartial response equalization ideal value at a sample time other thanthe top sample time and the last sample time, it becomes possible todetect the phase error at satisfactory accuracy.

Also, it is preferable for the phase error detecting device describedabove that the phase detection portion does not output the phase errorto the synchronization detection portion in a case where the phase errorthat has been detected is larger than a predetermined threshold value.

According to this configuration, the phase error is not outputted to thesynchronization detection portion in a case where the detected phaseerror is larger than the predetermined threshold value. It thus becomespossible to delete a phase error larger than the predetermined thresholdvalue, which can cause a disturbance. Hence, a reproduction clock signalcan be generated in a stable manner.

Also, it is preferable for the phase error detecting device describedabove that the waveform shaping portion includes a partial responseequalization filter that equalizes the digital reproduced signal and anadaptive coefficient update portion that adaptively updates coefficientsof the partial response equalization filter to have a desired partialresponse characteristic, and that the adaptive coefficient updateportion updates the coefficients of the partial response equalizationfilter to be right-left symmetrical.

Also, it is preferable for the phase error detecting device describedabove that the adaptive coefficient update portion includes an errorsignal detection portion that generates an equalization error signal toupdate the coefficients of the partial response equalization filter, andthat the error signal detection portion does not output the equalizationerror signal relating to a shortest mark or the equalization errorsignal equal to or larger than a predetermined value.

Also, it is preferable for the phase error detecting device describedabove that the adaptive coefficient update portion adaptively switchesan LMS (The Least-Mean Square) algorithm and a frequency samplingalgorithm in response to a state of the synchronization detectionportion, so that the adaptive coefficient update portion updates thecoefficients of the partial response equalization filter using one ofthe LMS algorithm and the frequency sampling algorithm.

Also, it is preferable for the phase error detecting device describedabove that the state of the synchronization detection portion includes astate in which frequency entrainment is controlled and a state in whichphase entrainment is controlled in a process to bring the digitalreproduced signal into synchronization with the reproduction clocksignal.

Also, it is preferable for the phase error detecting device describedabove that the maximum likelihood decoding is a method having at leasttwo pieces of edge information in state transition patterns having asmallest different metric among the differential metrics at a pluralityof merging points at which a set of paths branched from a given statemerges.

Also, it is preferable for the phase error detecting device describedabove that the phase detection portion detects the phase error byexcluding state transition patterns that contain a shortest mark.

Also, it is preferable for the phase error detecting device describedabove that the phase detection portion changes the state transitionpatters used to detect the phase error in response to the state of thesynchronization detection portion.

Also, it is preferable for the phase error detecting device describedabove that the state of the synchronization detection portion includes astate in which frequency entrainment is controlled and a state in whichphase entrainment is controlled in a process to bring the digitalreproduced signal into synchronization with the reproduction clocksignal.

Also, it is preferable for the phase error detecting device describedabove that the synchronization detection portion is furnished with acapability of appending a predetermined gain to the phase error detectedby the phase detection portion, normalizes the phase error with an idealdifferential metric distance in a process to extract the phase error inthe phase detection portion, and maintains the gain in a synchronizationloop almost constant by lowering the gain when scattering of the phaseerror is large and by increasing the gain when the scattering of thephase error is small.

Also, it is preferable for the phase error detecting device describedabove that the synchronization detection portion is furnished with acapability of adaptively changing state transition patterns used todetect the phase error in the phase detection portion, normalizes thephase error with an ideal differential metric distance in a process toextract the phase information in the phase detection portion, andmaintains the gain in a synchronization loop almost constant by making achange so as not to detect the phase error from state transition pattersrelating to a short mark when scattering of the phase error is large andby making a change so as to detect the phase error from the statetransition patterns relating to the short mark when the scattering ofthe phase error is small.

A phase error detection method according to another aspect of theinvention includes: a shaping step of receiving a digital reproducedsignal generated from an analog reproduced signal reproduced from aninformation recording medium and shaping a waveform of the digitalreproduced signal; a maximum likelihood decoding step of applyingmaximum likelihood decoding to the digital reproduced signal in thewaveform shaped in the shaping step and generating a binarized signalindicating a result of the maximum likelihood decoding; a phasedetecting step of detecting a phase error on the basis of the digitalreproduced signal in the waveform shaped in the shaping step and thebinarized signal generated in the maximum likelihood decoding step; anda synchronization detecting step of making synchronization with thereproduced signal using the phase error detected in the phase detectingstep. In the phase detecting step, the phase error is extracted duringthe maximum likelihood decoding using state transition patterns havingonly a single zero cross point among differential metrics at a pluralityof merging points at which a set of paths branched from a given statemerges.

An optical disc device according still another aspect of the inventionincludes an optical head and the phase error detecting device describedabove. According to this configuration, it is possible to apply thephase error detecting device described above to an optical disc.

An optical disc device according to still another aspect of theinvention includes: an A/D portion that generates s digital reproducedsignal from an analog reproduced signal reproduced from an informationrecording medium; an AGC/OFFSET control portion that controls anamplitude and an offset so as to fall within a predetermined D range ofthe A/D portion; a waveform shaping portion that shapes a waveform ofthe digital reproduced signal generated by the A/D portion; a maximumlikelihood decoding portion that applies maximum likelihood decoding tothe digital reproduced signal in the waveform shaped by the waveformshaping portion and generates a binarized signal indicating a result ofthe maximum likelihood decoding; a phase detection portion that detectsa phase error on the basis of the digital reproduced signal in thewaveform shaped by the waveform shaping portion and the binarized signalgenerated by the maximum likelihood decoding portion; and asynchronization detection portion that generates a reproduction clocksignal using the phase error detected by the phase detection portion andbrings the digital reproduced signal into synchronization with thereproduction clock signal that has been generated. The waveform shapingportion includes: a partial response equalization filter that equalizesthe digital reproduced signal; an adaptive coefficient update portionthat adaptively updates coefficients of the partial responseequalization filter so as to achieve a desired partial responsecharacteristic; and synchronization entrainment control portion thatcontrols an entrainment procedure for synchronization with the digitalreproduced signal. The synchronization entrainment control portion doesnot activate the adaptive coefficient update portion or the phasedetection portion until the AGC/OFFSET control portion converges withina predetermined range.

A waveform shaping device according to still another aspect of theinvention includes: a finite impulse response filter that receives adigital reproduced signal generated from an analog reproduced signalreproduced from an information recording medium and shapes a waveform ofthe digital reproduced signal; an amplitude level detection portion thatdetects an N′th amplitude level from a predetermined reference level inan output of the finite impulse response filter; an amplitude errordetection portion that detects a difference between the amplitude leveldetected by the amplitude level detection portion and a predeterminedtarget amplitude; and a coefficient calculation portion that calculatescoefficients of the finite impulse response filter on the basis of anoutput of the amplitude error detection portion. The coefficientcalculation portion calculates the coefficients of the finite impulseresponse filter so that the output of the amplitude error detectionportion reaches the predetermined target amplitude and updates thecoefficients of the finite impulse response filter to the coefficientsthat have been calculated.

According to this configuration, a digital reproduced signal generatedfrom an analog reproduced signal reproduced from an informationrecording medium is received and the waveform of the digital reproducedsignal is shaped by the finite impulse response filter. The N'thamplitude level from the predetermined reference level in an output ofthe finite impulse response filter is detected by the amplitude leveldetection portion. Subsequently, a difference between the detectedamplitude level and the predetermined target amplitude is detected bythe amplitude error detection portion, and the coefficients of thefinite impulse filter are calculated on the basis of an output of theamplitude error detection portion by the coefficient calculationportion. In this instance, the coefficients of the finite impulseresponse filter are calculated so that an output of the amplitude errordetection portion reaches the predetermined target amplitude and thecoefficients of the finite impulse response filter are updated to thecoefficients that have been calculated by the coefficient calculationportion.

It thus becomes possible to shape a digital reproduced signal generatedfrom an analog reproduced signal reproduced from an informationrecording medium into a waveform suitably used in the maximum likelihooddecoding. Accuracy in detecting the phase error can be thereforeenhanced, which makes it possible to generate a reproduction clocksignal in a stable manner.

Also, it is preferable for the waveform shaping device described abovethat: the amplitude level detection portion includes a first amplitudelevel detection portion that detects an N′th amplitude level from apredetermined reference level in an output of the finite impulseresponse filter, and a second amplitude level detection portion thatdetects an M′th (M>N) amplitude level from the predetermined referencelevel in the output of the finite impulse response filter; the amplitudeerror detection portion includes a first amplitude error detectionportion that detects a difference between the amplitude level detectedby the first amplitude level detection portion and a first targetamplitude, and a second amplitude error detection portion that detects adifference between the amplitude level detected by the second amplitudelevel detection portion and a second target amplitude different from thefirst target amplitude; and the coefficient calculation portioncalculates the coefficients of the finite impulse response filter on thebasis of outputs of the first amplitude error detection portion and thesecond amplitude error detection portion.

According to this configuration, the N′th amplitude level from thepredetermined reference level in an output of the finite impulseresponse filter is detected by the first amplitude level detectionportion. The M'th (M>N) amplitude level from the predetermined referencelevel in an output of the finite impulse response filter is detected bythe second amplitude level detection portion. A difference between theamplitude level detected by the first amplitude level detection portionand the first target amplitude is detected by the first amplitude errordetection portion. A difference between the amplitude level detected bythe second amplitude level detection portion and the second targetamplitude different from the first target amplitude is detected by thesecond amplitude error detection portion. Coefficients of the finiteimpulse response filter are calculated on the basis of the outputs ofthe first amplitude error detection portion and the second amplitudeerror detection portion by the coefficient calculation portion.

Hence, not only can a difference between the N′th amplitude level fromthe predetermined reference level in an output of the finite impulseresponse filter and the first target amplitude be detected, but also adifference between the M′th (M>N) amplitude level from the predeterminedreference level in an output of the finite impulse response filter andthe second target amplitude can be detected. Because the coefficients ofthe finite impulse response filter are calculated on the basis of theseamplitude errors that have been detected, it becomes possible tocalculate the coefficients of the finite impulse response filter in amore reliable manner.

Also, it is preferable for the waveform shaping device described abovethat the coefficient calculation portion updates the coefficients of thefinite impulse response filter using a frequency sampling algorithm.

Also, it is preferable for the waveform shaping device described abovethat the finite impulse response filter is a digital filter holdingcoefficients of nine taps and controls five gains in predeterminedfrequency bands using the frequency sampling algorithm in such a mannerthat a gain in a lowest band is a gain of one, a gain in a second lowestband is controlled so that an output of the second amplitude errordetection portion becomes smaller, a gain in a third lowest band iscontrolled so that an output of the first amplitude error detectionportion becomes smaller, a gain in a fourth lowest band is controlled sothat the output of the first amplitude error detection portion becomessmaller, and a gain in a fifth lowest band is a fixed gain of one orless.

Also, it is preferable for the waveform shaping device described abovethat the finite impulse response filter is a digital filter holdingcoefficients of seven taps and controls four gains in predeterminedfrequency bands using the frequency sampling algorithm in such a mannerthat a gain in a lowest band is a gain of one, a gain in a second lowestband is controlled so that an output of the second amplitude errordetection portion becomes smaller, a gain in a third lowest band iscontrolled so that an output of the first amplitude error detectionportion becomes smaller, and a gain in a fourth lowest band is a fixedgain of one or less.

A waveform shaping method according to still another aspect of theinvention includes: a waveform shaping step of receiving a digitalreproduced signal generated from an analog reproduced signal reproducedfrom an information recording medium and shaping a waveform of thedigital reproduced signal by allowing the digital reproduced signal topass through a finite impulse response filter; an amplitude leveldetecting step of detecting an N′th amplitude level from a predeterminedreference level in an output of the finite impulse response filter; anamplitude error detecting step of detecting a difference between theamplitude level detected in the amplitude level detecting step and apredetermined target amplitude; and a coefficient calculating step ofcalculating coefficients of the finite impulse response filter on thebasis of an output in the amplitude error detecting step. In thecoefficient calculating step, the coefficients of the finite impulseresponse filter are calculated so that the output in the amplitude errordetecting step reaches the predetermined target amplitude and thecoefficients of the finite impulse response filter are updated to thecoefficients that have been calculated.

An optical disc device according to still another aspect of theinvention includes an optical head, and the waveform shaping devicedescribed above. According to this configuration, it is possible toapply the waveform shaping device described above to an optical disc.

The phase error detecting device, the waveform shaping device, and theoptical disc device of the invention are able to generate a reproductionclock signal in a stable manner, and are therefore useful as a phaseerror detecting device, a waveform shaping device, and an optical discdevice that perform signal processing using the maximum likelihooddecoding method.

This application is based on Japanese Patent Application No. 2007-339921filed on Dec. 28, 2007, the contents of which are hereby incorporated byreference.

It should be appreciated that specific embodiments or examples describedin the detailed description of the invention are intended merely to makethe technical contents of the invention clear. The invention thereforeshould not be understood in a narrow scope limited by such specificexamples and the invention can be modified in various manners within thespirit of the invention and the scope of appended claims.

1. A phase error detecting device, comprising: a waveform shapingportion that receives a digital reproduced signal generated from ananalog reproduced signal reproduced from an information recording mediumand shapes a waveform of the digital reproduced signal; a maximumlikelihood decoding portion that applies maximum likelihood decoding tothe digital reproduced signal in the waveform shaped by the waveformshaping portion and generates a binarized signal indicating a result ofthe maximum likelihood decoding; a phase detection portion that detectsa phase error on the basis of the digital reproduced signal in thewaveform shaped by the waveform shaping portion and the binarized signalgenerated by the maximum likelihood decoding portion; and asynchronization detection portion that generates a reproduction clocksignal using the phase error detected by the phase detection portion andbrings the digital reproduced signal in synchronization with thereproduction clock signal that has been generated, wherein the phasedetection portion extracts, during the maximum likelihood decoding, thephase error using state transition patterns having only a single zerocross point among differential metrics at a plurality of merging pointsat which a set of paths branched from a given state merges.
 2. The phaseerror detecting device according to claim 1, wherein: a relation betweena spatial frequency and a signal amplitude is defined by an opticaltransfer function, the signal amplitude decreases almost linearly as thespatial frequency becomes higher, and a limit of reproduction at whichthe signal amplitude decreases to 0 is defined as a cutoff frequency ofthe optical transfer function, then a shortest mark frequency of areproduced signal recorded in the information recording medium is in avicinity of the cutoff frequency of the optical transfer function. 3.The phase error detecting device according to claim 1, wherein: thephase detection portion extracts, during the maximum likelihooddecoding, the phase error using a partial response equalization idealvalue at a second sample time at which influences of distortion of awaveform of an input signal are less than at a first sample time fromthe state transition patterns having only a single zero cross pointamong the differential metrics at a plurality of merging points at whicha set of paths branched from a given state merges.
 4. The phase errordetecting device according to claim 3, wherein: the first sample-timeincludes a top sample time and a last sample time; the second sampletime includes a sample time other than the top sample time and the lastsample time; and the phase detection portion extracts, during themaximum likelihood decoding, the phase error using the partial responseequalization ideal value at a sample time other than the top sample timeand the last sample time from the state transition patterns having onlya single zero cross point among the differential metrics at a pluralityof merging points at which a set of paths branched from a given statemerges.
 5. The phase error detecting device according to claim 1,wherein: the phase detection portion does not output the phase error tothe synchronization detection portion in a case where the phase errorthat has been detected is larger than a predetermined threshold value.6. An optical disc device, comprising: an optical head; and the phaseerror detecting device set forth in claim
 1. 7. A waveform shapingdevice, comprising: a finite impulse response filter that receives adigital reproduced signal generated from an analog reproduced signalreproduced from an information recording medium and shapes a waveform ofthe digital reproduced signal; an amplitude level detection portion thatdetects an N′ th amplitude level from a predetermined reference level inan output of the finite impulse response filter; an amplitude errordetection portion that detects a difference between the amplitude leveldetected by the amplitude level detection portion and a predeterminedtarget amplitude; and a coefficient calculation portion that calculatescoefficients of the finite impulse response filter on the basis of anoutput of the amplitude error detection portion, wherein the coefficientcalculation portion calculates the coefficients of the finite impulseresponse filter so that the output of the amplitude error detectionportion reaches the predetermined target amplitude and updates thecoefficients of the finite impulse response filter to the coefficientsthat have been calculated.
 8. The waveform shaping device according toclaim 7, wherein the amplitude level detection portion includes: a firstamplitude level detection portion that detects an N′th amplitude levelfrom the predetermined reference level in the output of the finiteimpulse response filter; and a second amplitude level detection portionthat detects an M'th (M>N) amplitude level from the predeterminedreference level in the output of the finite impulse response filter,wherein the amplitude error detection portion includes: a firstamplitude error detection portion that detects a difference between theamplitude level detected by the first amplitude level detection portionand a first target amplitude; and a second amplitude error detectionportion that detects a difference between the amplitude level detectedby the second amplitude level detection portion and a second targetamplitude different from the first target amplitude, and wherein thecoefficient calculation portion calculates the coefficients of thefinite impulse response filter on the basis of outputs of the firstamplitude error detection portion and the second amplitude errordetection portion.
 9. An optical disc device, comprising: an opticalhead; and the waveform shaping device set forth in claim 7.